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WIRELESS
SERVICING MANUAL
By
W. T. Cocking
m.iYe
MEMBER OF WIRELESS WORLD” TECHNICAL STAFF
Illustrated with 1 30 Diagrams and 22 Photographs
SEVENTH EDITION
(Revised)
Published for
“ Wireless VVorld ”
by
ILIFFE & SONS LIE
LONDON. iIRMINGHAM, COVINTRY, MANCHKTBR AND OLAIQOW
Published for " Wireless World ’* by lliffe and Sons, Ltd., Dorset House,
Stafn ford Street, London, S.E.I.
First Published
jjLMay. 1936
Second Edition
December, 1936
Third Edition
1937
Fourth Edition
1938
Fifth Edition
1940
Sixth Edition
May, 1941
Second Impression
August, 1941
Third Impression
January, 1942
Fourth Impression
June, 1942
Fifth Impression
February, 1943
Sixth Impression
August, 1943
Seventh Impression
January, 1944
Eighth Impression
December, 1944
Seventh Edition
January, 1946
Second Impression (Revised)
October, 1946
Third Impression
1948
This book is produced in complete conformity
with the authorized economy standards
Made and printed in Great Britain by The Chapel
River Press, Andover, Hants
[Bks. 228/3.48]
CONTENTS
CHAPTER PAGE
Preface vii
1 Testing Equipment i
2 Current and Voltage Testing 9
3 The Interpretation of Meter Readings i6
4 Valves 25
5 Tracing Mains Hum 29
6 Motor-Boating 47
7 Instability in R.F. and I.F. Stages 55
8 Frequency and Amplitude Distortion 65
9 Background Noise and Local Interference 81
10 The Adjustment of Ganging — Straight Sets 91
11 The Adjustment of Ganging — Superheterodyne
I.F. Amplifiers 104
12 The Adjustment of Ganging — Superheterodyne
Signal-Frequency Circuits 115
13 Tracing the Cause of Whistles 129
14 Superheterodyne Whistles 136
15 Frequency-Changers 148
r6 Automatic Volume Control Systems 169
17 Short-Wave Receivers 180
18 Short-Wave Converters 193
19 The Loud Speaker 199
20 Extension Loud Speakers 205
21 The Aerial-Earth System 209
22 Automatic Frequency Control 217
23 Push-Pull Amplifiers 228
24 Miscellaneous Defects and Adjustments 244
25 Negative Feed-Back 250
26 Television Receivers 257
27 The Use of Cathode-Ray Test Gear 270
APPENDIX
I Common Circuit Constants 277
Standard Symbols 279
Useful Formul.^ 280
Superheterodyne Ganging 282
Short-Wave Aerials and Feeders 285
Copper Wire Tables 286
II Colour Codes 288
III Capacity and Resistance Bridge 292
IV Valve Testing Bridge 301
V Valve Data 309
Index 324
V
LIST OF HALF-TONE ILLUSTRATIONS
PAGE
Avometer, Model 7, and D.C. Avominor 2
Muirhead Precision Resistors and Capacitors 4
Sangamo-Weston Analyser 6
Wireless World Capacity and Resistance Bridge 7
Ferranti A.C, Test Set 15
Taylor Valve Tester 27
Avo Valve Characteristic Meter 28
British Physical Laboratories Test Sbt 54
Belling-Lee Interference Suppressor 86
Taylor Signal Generator 103
British Physical Laboratories’ Signal Generator 113
Avo All-Wave Oscillator 113
Advance Components Attenuator x 14
Labgear C.R. Tube Viewing Unit 128
Taylor Test Meter 135
Ultra-Short- Wave Converter 194
Hunts Capacitance and Resistance Analyser 269
Wireless World Capacity and Resistance Bridge Details
293, 294, 299
Wireless World Valve Testing Bridge 302, 305
VI
PREFACE
B roadcasting has now such a definite place in the
daily lives of all sections of the community, and is so
greatly relied upon for entertainment and instruction,
that it is impossible to contemplate with equanimity any break-
down in the system. A general cessation of broadcasting is
fortunately an event of extreme improbability, but a breakdown
in a receiver is by no means uncommon and to its owner it is
just as disastrous.
The repairing of receiving equipment has consequently
become an important section of the wireless industry. The
growing standard of performance which is demanded of
receivers, and their resulting complexity, necessitate a higher
degree of skill on the part of the repairer than ever before, and
a degree of skill which will certainly not grow less with the
passage of time. Radio Servicing is thus becoming a highly-
specialized profession, and the days of the self-styled “ experts ”
with no real knowledge of radio technique are for ever gone.
The location of a defect in a commercially-produced receiver,
the initial adjustments of a home-constructed set, and the
testing of experimental apparatus all call for the application of
service principles. Familiarity with these principles is not
alone sufficient, however, and the proper testing gear is also
necessary, together with an adequate understanding of wireless
phenomena. Speed and certainty in the diagnosis and treat-
ment of receiver ailments come only with experience. Methods
can be taught, however, and to approach a problem correctly is
half way to the solution.
The immediate concern of this book is with purely servicing
problems — with the location and cure of the hundred and one
defects which can develop in a receiver and its associated
equipment. The choice of suitable testing gear is briefly
discussed, however, for this is a matter of considerable im-
portance upon which it is not always easy to obtain unbiased
information. No attempt has been made to treat wireless
theory or to explain the mode of operation, save in one or two
isolated cases where the circuit is so new that an adequate
description is unlikely to be found in any standard textbook.
Those who lack the necessary understanding of basic prin-
ciples are recommended to acquire it from a good textbook,
and even with the most elementary knowledge of mathematics
it is possible to gain sufficient acquaintance with theory for all
vii
PREFACE
service matters. Although mathematical ability may be
essential to the designer, it is not to the service engineer.
Excellent as they may be, however, textbooks are not sufficient
to enable him to keep up-to-date, for in wireless, invention
succeeds invention so rapidly that it is possible to remain
au Jait with current developments only by the regular perusal
of one or more reliable technical journals.
In compiling this book, the author’s aim has been to describe
those methods which he has himself found to be the most
suitable for the location of defects. The arrangement of
material, however, is one of “ symptoms ” rather than
“ defects ”, since he believes that this makes the book handier
for reference ; and with this same end in view particular
attention has been paid to the index. The appendix contains
in tabular form much information commonly required by
service engineers, including valve base connections for British,
American and Continental valves. Here every endeavour has
been made to include obsolete as well as current types, for
receivers embodying old types of valves are still met with
occasionally and information as to the connections is not
always readily available.
W. T. Cocking
Dorset House^
Stamford Street,
London, S,E.i.
viil
WIRELESS SERVICING MANUAL
CHAPTER I
TESTING EQUIPMENT
R apid and efficient servicing demands the use of the
proper tools. They are neither many nor unduly
expensive, but they are essential to all who seek to place
their profession on a profitable basis. It is in many cases
quite possible for the expert to locate defects in receivers and
to repair them satisfactorily with but little apparatus ; yet
much time is wasted, and time is a very important factor in
any organization of servicing on a commercial basis.
The apparatus should be of good quality. Reliability and
ruggedness are of the first importance, and a laboratory standard
of accuracy is unnecessary. The service man should be able
to take D.C. measurements of voltage within ± 5 per cent,
from I volt to at least 600 volts, and current with the same
accuracy from 0*5 mA to i ampere ; for A.C. work, voltages
between 2 volts and 1,000 volts will be encountered, but current
measurements are not often required. He must be able to
check the values of resistances to within ± 10 per cent, for
values between i ohm and at least 2 megohms, and the values
of condensers between 50 and 4 juF. He must have a
modulated test oscillator and output meter, a set of valve
adaptors and such tools as pliers, screwdrivers, wirecutters,
soldering-iron, and so on.
Although a range of different voltmeters and milliammeters
is extremely useful and enables different measurements to be
made simultaneously, a multi-range A.C. and D.C. voltmeter
and milliammeter, which may also be combined with an
ohmmeter, is just as useful for most service work. Instruments
of this nature are available commercially at reasonable prices,
but when selecting one care should be taken to see that the
resistance of the voltmeter is high. It is, however, quite easy
to construct such an instrument around a single low-range
milliammeter. If this course be adopted, a moving-coil milli-
ammeter of good quality should be obtained, and it will pay to
buy as good an instrument as one can afford. The diameter
of the scale should not be less than some 2J in., and a meter
having a full-scale deflection for a current of i mA is best.
The connection of a suitable resistance in series with such a
meter will convert it into a voltmeter, and since the meter
I
WlREl^SS SERVICING MANUAL
resistance is negligible compared with the added resistance for
all ranges normally required, the necessary value of added
resistance in ohms equals the voltage required for full-scale
deflection multiplied by i,ooo.
The connections for a three-range voltmeter with resistance
values are shown in Fig. 2 : such an instrument has ranges of
o-io, 0-100, and 0-1,000 volts with an internal resistance of
1,000 ohms per volt. If accurate resistances are used, no
calibration is needed, and it is only necessary to multiply the
meter reading by 10 for the o-io volts range, by 100 for the
0-100 volts range, and by 1,000 for the 0-1,000 volts range to
obtain the correct voltage. For most purposes it will suffice to
buy wire-wound resistances accurate within i i per cent., and
these can be obtained quite reasonably, for most makers can
supply resistances of this accuracy for a small extra charge.
TESTING EQUIPMENT
A power rating of i watt
is adequate, and the
900,000 ohms resistance
should be made up of
several of lower value
connected in series.
A multi-range milli-
ammeter can be built
with the same basic
instrument in several
different ways. One of
the best is shown in
Fig. 3. The meter is
shunted by a resistance
having a value such that
the meter reads full-scale for a current of 10 mA ; tapping
points are then selected so that full-scale deflections for
1 00 mA and i,ooo mA (i ampere) are obtained. The
total value of the shunt resistance should be one-ninth of the
meter resistance. This is usually about 100 ohms, so that
the shunt should be about 1 1 ohms, but as the meter resistance
is rarely accurately known, the value of the shunt must be
determined experimentally, as also must the tapping points,
with the aid of another meter.
Th(^ subject of making such gear is not one which can be
entered into here, but it may be said that it is quite possible
to make an A.C.
voltmeter and milliam-
meter out of the
same basic meter by
adding a suitable metal
rectifier. The whole
apparatus can be
combined in one case
with switches to make
the necessary circuit
changes and a multi-
range D.C. and A.C.
voltmeter and milliam-
meter obtained.
Since the response is
usually good up to 10
kc/s it can also be used
as an output meter.
Fig. 2 : This diagram shows the connections of a
typical multi-range voltmeter
WIRELESS SERVICING MANUAL
The Ohmmeter
The ohmmeter is equally easy to deal with, for basically it is
nothing more than a voltmeter and suitable battery. Suppose,
for instance, that the voltmeter (resistance i,ooo ohms per volt)
is set on the lo-volts range and a battery of exactly lo volts is
connected in series with one of its leads. Save that it is un-
calibrated, the gear is then an ohmmeter. It will read full-scale
if the external circuit has zero resistance and read nothing if it
has infinite resistance. When the circuit has the same resistance
as the meter, io,ooo ohms, the meter will read half-scale.
Calibration for different resistance values is readily obtained
either by calculation or by the insertion of known values. In
practice, however, ohmmeters are somewhat more complex, but
only because means are incorporated for compensating for
variations in the battery voltage.
Although the ohmmeter is an exceedingly useful piece of
equipment it is rarely very accurate, for even when correction
is applied the falling* battery voltage introduces some error, and
the scale is an awkward one, being very cramped for high
resistances. The ohmmeter thus finds its greatest application
in continuity testing and in the rough measurement of
resistances. For more accurate work a bridge is advisable.
In general testing it is exceedingly important to have means
of checking the capacity of condensers, for components having
an internal open-circuit or wrongly-marked values are by no
means unknown. With an A.C. meter it is possible to devise
TESTING EQUIPMENT
a method of measuring capacity which is similar to the ohm-
meter for resistance. It is not a very satisfactory method,
however, for it necessitates an A.C. supply, which may not
always be available, and it is only reasonably accurate with
large capacity condensers. The bridge method is far better.
Inexpensive bridges for resistance and capacity measurements
are commercially available, and constructional details of a
simple bridge are given in Appendix III. This instrument
will give direct readings of both resistance and capacity over
the wide range ofio/A/iFtoio/xF and lo ohms to lo megohms.
It consists of two parts — the bridge proper and an A.F. oscillator
for exciting it. The latter has self-contained batteries and may
also be used for modulating an R.F. oscillator when required.
The measurement of inductance is much more difficult. It
is rarely required to check A.F. inductances in servicing, but
it would undoubtedly be useful to have a method of measuring
the inductance of tuning coils. Unfortunately, however,
accurate methods are essential for these and measurements are
of little value unless they can be relied upon to at least zt i per
cent. A good inductance bridge is much more expensive than
the equipment which will serve for capacity tests and it is
hardly portable. For the service shop, where many tests have
to be carried out, however, the Baldwin Inductance Bridge can
be recommended.
Essential Test Oscillator
Although an inductance bridge may be something of a
luxury, a test oscillator is almost essential. It should cover
the range of 0*1-25 Mc/s at least and be accurately calibrated,
particularly over the band of 420—500 kc/s, within which the
intermediate frequencies of most superheterodynes lie. It
should be well screened and its output should be adjustable
over a wide range — 100 fiv to i v, and the minimum is better
10 fJLV or less. It should be modulated and arranged so that
the modulation can be switched on and off at will. The type
of oscillator which is fitted with separate coils is usually to be
preferred to that which relies upon harmonics for a wide
tuning range.
The test oscillator is used mainly to provide a known signal
for ganging purposes, and quite a simple one can be satisfactory.
The most important points to look for are the accuracy of the
frequency calibration, an adequate range of output, and an
absence of reaction of the output control upon frequency.
This last point is particularly important and one in which the
5
WIRELESS SERVICING MANUAL
cheapest oscillators often fail badly. Its importance will be
realised when it is remembered that in ganging a set it is usually
essential for a whole series of adjustments to be made at a
single frequency. As the adjustments arc carried out the gain
of the set varies, and the output of the test oscillator has to be
altered to suit it. Clearly, then, it will not be satisfactory if
this alteration of the output changes the frequency.
One must, however, be realistic and recognise that complete
freedom from the effect is unlikely to be found in anything but
expensive equipment. In practice, it will normally suffice if
the frequency does not change by more than i part in i,ooo
for the whole range of output. For a change in output below
I mV., the change of frequency should not exceed i part in
10,000. Fortunately, in practice, the greatest alteration of
TESTING EQUIPMENT
frequency is usually found for changes of output near the
maximum .
For laboratory and experimental work, as distinct from
repairing, a standard signal generator is much better than a
test oscillator, but is much more expensive. The two instru-
ments are essentially similar and the signal generator is really a
highly refined test oscillator. It is much better screened and
it contains meters which enable a known depth of modulation
to be used and a known R.F. output to be applied to the
attentuator. The attentuator is accurately made and gives a
known attenuation.
The signal generator, therefore, is essentially a measuring
instrument providing a modulated R.F. output of known
amplitude, frequency and depth of modulation. The test
oscillator gives the same kind of output but neither the R.F.
output nor the depth of modulation are known with any degree
of accuracy. Of course, there are all grades of both instruments,
and a low grade signal generator may be almost indistinguishable
from a first-class test oscillator.
For use with the test oscillator an output meter is very
desirable, but not essential. It consists basically of an A.C.
voltmeter of the rectifier type with a transformer. Some
instruments have a tapped transformer so that the meter
impedance can be matched to the valve, but a fixed ratio is
7
WIRELESS SERVICING MANUAL
Fig, 6 : The basic voltage supply arrangements are shown in outline, coupling components
and by-pass condensers being omitted. At (o) cathode bias is used, while at (b) a
resistance in the negative H.T. lead supplies the bias ; at (c) a battery is used
useless to attempt to measure the grid bias at the grid of the
valve, and it is in fact dangerous from the point of view of
valve life to attempt to do so. The resistance of the grid
circuit of most valves is so high that the connection of any
ordinary meter to the valve grid will reduce the bias actually
applied to the valve to perhaps one-tenth of its normal value,
and it is obvious that this is likely to prove harmful to the
valve. The grid bias, therefore, must be measured at its
source and not on the valve. In Fig. 6 (c) the voltage across the
bias battery should be checked ; in (b) the voltage across the
bias resistance must be measured with the voltmeter joined
between negative H.T. and the chassis.
Cathode-Bias Resistances
When dealing with receivers in which cathode-bias re-
sistances are used (a ) , and they are probably the majority, the
negative terminal of the meter can be left connected to the
chassis, for all voltages are positive with respect to this point.
The grid bias is now the voltage between chassis and cathode,
fO
CURRENT AND VOLTAGE TESTING
and although it is usually listed as the grid voltage it is much
better and clearer to call it the cathode voltage.
The voltage measured between anode and chassis is easily
seen to be not the true anode voltage but the sum of the anode
and cathode voltages. The difference is usually insignificant,
except in the case of output valves, for the anode voltage is
usually 150-250 volts and the cathode voltage is rarely greater
'than 0*5-3 volts, unless a manual volume control operating on
a variable-mu valve is used. Apart from these exceptions, the
difference between the true anode voltage and that measured
from the chassis is negligibly small, and is often less than the
permissible error in the meter.
In the case of many output valves, however, it is important
to distinguish between the two figures. With a PX4 valve, for
instance, the true anode voltage might be 250 volts and the
grid bias — 33 volts. Taking all measurements from the
chassis, therefore, the cathode (filament in this case) would be
■i" 33 volts and the anode + 283 volts. The difference is even
greater with valves of the LS6A type, for the grid bias is — 93
volts for an anode potential of 400 volts, and it is obvious that
much confusion may arise if care is not taken to distinguish
between the two methods of measurement.
Similarity with a Variable-Mu Valve
A similar case arises with a variable-mu valve having a
manual volume control of the types shown in Fig. 7. Volume
is controlled by varying the grid bias applied to the valve, and
in each case it is really the cathode voltage which changes.
At (a) the cathode voltage is a minimum when the slider of the
potentiometer is nearest the earth line, and at (6) when the
variable resistance is a minimum. Measurement will then
indicate normal voltages, perhaps 200 for the anode and 80 for
the screen grid. When the volume control is set for weakest
volume, however, the apparent potential of the anode may be
250-300 volts and that of the screen grid 120 volts or so. The
cathode, however, may be 30-40 volts positive with respect to
the chassis so that the true anode and screen voltages are the
apparent values less the cathode voltage, and the rise in voltage
is much less marked than it appears.
It should, of course, always be remembered that no voltmeter
of the current-consuming type ever indicates exactly the voltage
existing in the circuit when the voltmeter is absent ; it only
shows the voltage existing while it is connected. This is due
to the fact that the voltmeter itself consumes a certain amount
WIRELESS SERVICING MANUAL
of power which must be supplied by the circuit to which it is
connected. In order that the readings may be reasonably
accurate in circuits not having a good voltage regulation it is
necessary, therefore, for the resistance of the meter to be high
compared with that of the circuit across which it is connected.
If the voltage at a point of poor regulation — the anode
voltage of a resistance-coupled valve, for instance — is measured
by two different voltmeters of different resistances, the readings
will not be the same. The lower resistance meter will show
the lower voltage. It will be understood, of course, that if
both meters were simultaneously connected to the circuit, they
would read alike, but their indication would be less than that
given by either alone.
It will be clear therefore, that, to be of certain help in
diagnosing faults in a receiver, the voltage readings must be
properly interpreted. Absolute reliance can be placed upon
them only when there is knowledge of the indications which
should be given by the particular meter used. In spite of this,
they are of great value, for the errors introduced by the meter
CURRENT AND VOLTAGE TESTING
in taking measurements in the
average receiver are only likely
to be misleading at certain points,
since the errors introduced are
unlikely to exceed about lo per
cent, and variations between
components and valves may
easily cause such a discrepancy
without seriously affecting the
performance. The serious errors
occur only in circuits of poor
regulation and it is then easy to
check in a different manner.
Using the Milliammeter
Current measurements do not
suffer from the same tendency
to inaccuracy, for, although the
meter must consume some
power, in practically every wire-
less circuit its resistance is
negligible in comparsion with
the circuit resistance. Current
measurements, therefore, can be
relied upon absolutely, the sole
limitation being the accuracy
practice, the only precautions which it is necessary to take are
to see that no current flows through the meter other than that
which it is desired to measure, and to make sure that its intro-
duction to the circuit does not cause instability.
The milliammeter must be included in series with the circuit,
so that it is somewhat more troublesome to use than a voltmeter.
Referring to Fig. 8, the meter can be inserted at any convenient
point between positive H.T. and the anode of the valve, and at
whichever point it be inserted the reading will be the same.
Where possible, however, it is wise to include the meter between
the decoupling resistance Ri and the H.T. line, for although
the current is the same in all parts of a series circuit, the leads
to the meter may introduce instability in certain circuits if the
meter be included at other points. Instability, of course, is
likely to cause a big change in the anode current of the valve
and so render the reading useless.
In general testing, however, it is often inconvenient to
include the meter in the theoretically correct position, for it
13
A.C. potential, generally between the
decoupling resistance RI and the H.T.
line
of the meter itself. In
WIRELESS SERVICING MANUAL
involves the identification
of the decoupling resist-
ance, the unsoldering of
the lead to one end of it,
and the insertion of the
meter. With valves
having a top-cap anode
terminal, it is obviously
much simpler to insert
the meter between the
valve terminal and the
lead itself, while when the
anode is brought out to
a pin in the valve-base.
Fig. 9 : A splic-stnode valve adaptor is a valuable the USe of a Split-anodc
asset to the measurement of anode current, and valve adaptor enables the
to prevent its promoting instability a condenser ^ , ,
should be wired between grid and catiiodo meter tO be COnneCteu
without disturbing in any
way the wiring of the set. It is consequently very easy and
speedy to take readings of anode and screen currents. It must
be remembered, however, that since the meter is connected
directly to the valve anode, its presence may introduce
instability.
No trouble is usually found in A.F. circuits, and a split-
anode adaptor can be used with impunity in all ordinary
apparatus. In the case of R.F. and I.F. stages, however, it is
another matter, and to prevent oscillation, it is a wise plan to
short-circuit the coil in the grid circuit of the valve. The
tuning condenser should not be short-circuited since in many
cases this would alter the grid bias on the valve. As an alter-
native to the short-circuit, a condenser of o-ooi /xF or so could
be connected between grid and chassis so that the input circuit
is widely mistuned and instability made most unlikely.
The most convenient course is to mount such a condenser
on the adaptor itself, wired between grid and cathode, so that
its insertion becomes automatic, and this is shown in Fig. 9.
Although the adaptor is unnecessary when taking anode current
readings with screen-grid valves having top-anode connections,
being then used only for screen current, it should still be
employed and the leads from it shorted in order that the
grid-cathode condenser may remain in circuit.
A five-pin adaptor will enable measurements to be taken
throughout the older mains sets and many battery sets ; in
some cases, however, a four-pin adaptor is needed for the
14
CURRENT AND VOLTAGE TESTING
latter, depending on whether the valveholders will permit the
passage of the centre-pin or not. Later receivers demand a
seven-pin adaptor in addition, and a few sets, a nine-pin
adaptor. Nowadays, however, octal-base valves are most
frequently used, so that quite a large range of adaptors is
needed if one is to be equipped to deal with all sets in this way.
Fig. 10 : The Ferranti A.C. Test Set : similar equipment is available for D.C.
measurements
15
CHAPTER 3
THE INTERPRETATION OF METER READINGS
T he great advantage of voltage and current testing is that
in many cases it leads one by a very short path to the
source of the trouble. The first step, of course, is to
check the voltages on all the valves of the receiver, and the
currents passed by their anodes and screen-grids. Often it
will be found that in one or two places the meter readings are
markedly different from the correct ones, and it may then be
possible to cut short the procedure and investigate the circuits
of the valve concerned straight away.
In general, however, the figures should be tabulated, and it
is convenient to have a standard set of forms for this purpose.
In this way full data on the receiver arc available for reference
at any time and are useful not only in the event of that particular
set developing a fault at a later date, but also when one has
another receiver of the same type for repair. Moreover, if one
has many sets of the same model to service and a careful record
be kept of the defects and periodically examined, the commonest
fault is soon revealed and often leads to a short-cut in in-
vestigating those models.
Having obtained a set of figures for a receiver, it is necessary
to interpret them correctly if they are to be of value. Suppose
that both voltages and currents are below normal throughout
the receiver, what does this mean ? It is at once obvious that
the most probable cause is a defective rectifier valve, if the set
be of the A.C. or A.C./D.C. type, or a run-down H.T. battery
with a battery set. In the latter case, however, the defect
should have been found earlier, for in testing a battery set the
first thing to do is always to check the batteries under load.
Reverting to the mains set, if the rectifier is not at fault, low
output can be due only to subnormal voltages on the secondaries
of the mains transformer, or to a low-resistance leak in one of
the smoothing condensers. The mains voltage will have been
checked during the general testing and if it be low may account
for the trouble. Do not forget to check that the mains are
connected to the correct tappings on the primary.
If the mains voltage is low, it is tempting to correct for it by
altering the connections to the transformer, but it is hardly safe
to do so. Low mains voltage usually occurs when the house is
a considerable distance from a sub-station, and is due to the
voltage drop in the cables. This voltage drop varies with the
16
THE INTERPRETATION OF METER READINGS
current and is likely to
be at its greatest on a
winter evening when
there is a big consump-
tion for lighting and
heating. In the daytime,
however, the load may
be small, and the voltage
drop negligible. The
voltage is then often
above the nominal
voltage, for the supply
at the sub-station is
usually somewhat higher
than the nominal voltage
in order to compensate
for the loss in the cables.
In cases of very bad
regulation, 240-volts
mains may deliver a
Fig. II: A typical circuit of mains equipment,
showing the points at which A.C. voltage
measurements should be taken
supply at as much as
260 volts in the daytime and as little as 215 volts at night. Such
cases are quite rare, however, and in general the fluctuations
are kept within the limits of ± 4 per cent. It can be seen that
the wide fluctuations sometimes found render it unsafe to
compensate for low voltage by altering the tapping on the mains
transformer primary, for abnormally low voltages usually mean
wide variations and a correspondingly high voltage at other
times .
The Mains Equipment
If the mains voltage be correct, the mains transformer
secondary voltage should be checked with an A.C. meter. The
total R.M.S. voltage across the H.T. secondary rarely exceeds
1,000 volts even with large output stages, and can consequently
be measured with a meter having a slightly higher full-scale
reading, such as 1,200 volts, by connecting it across the full
winding (points CD, Fig. ii). Care should be taken to use
thoroughly insulated test prods for the terminations to the
meter leads.
If the meter does not read high enough for this measurement,
each half of the winding can be checked separately by connecting
the meter first to CE and then to DE, and adding the figures
to obtain the full voltage. Ideally, the voltages across the two
17
WIRELESS SERVICING MANUAL
halves of the winding should be the same, but in practice they
may differ slightly.
The filament voltage of the rectifier should be measured with
the meter connected at AB, directly to the filament legs of the
valve-holder, in order to allow for the drop in the connecting
leads. * In modern sets it is usually either 4 or 5 volts according
to the type of rectifier valve, but in old sets voltages of 7*5 volts
and 5*5 volts were often used. Although the A.C. voltage is
small, do not forget that the filament of this valve is anything
up to 500 volts more positive than the chassis, and connect the
meter with care.
Now if the voltages throughout the receiver are abnormally
high, but the currents about normal or somewhat low, the
most probable cause is low heater voltage on the valves and
this should be checked by connecting an A.C. voltmeter to the
valve-heaters themselves. If the voltages are low, check the
output of the mains transformer (FG, Fig. ii) and if this is
correct the trouble obviously lies in the connecting leads.
When the transformer gives an output of 4 volts and the
valves require 4 volts, the connecting leads should not be of
thinner wire than No. 16 gauge or its equivalent in stranded
wire. Even with this gauge, an appreciable drop will occur if
there are many valves. In commercial sets, therefore, the
secondary voltage is often made slightly greater than 4 volts
to allow for the drop in the wiring. The lower current taken
by modern 6*3-volt valves has greatly eased matters, and
specially heavy conductors for the heater wiring are rarely
necessary with such valves. Incidentally, the accuracy of the
position of the centre-tap can be checked by measuring the
voltages across the two halves (FH,GH,Fig. ii) — they should
be equal.
When interpreting the figures for a receiver, the most im-
portant thing to remember is that a high anode voltage should
be accompanied by a high anode current if all is in order in
the particular circuit to which the figures apply. In the case
of pentodes and screen-grid valves, however, the anode current
is not greatly affected by the anode voljage, but is very dependent
on the screen potential, a high voltage at this point leading to
a marked increase in anode current.
Cathode Bias
When cathode bias is used, however, the increase in anode
current will not be proportional to the rise in anode voltage.
Thus, suppose that when investigating an output stage the
18
THE INTERPRETATION OF METER READINGS
valve is found to have an anode potential (anode-cathode) of
300 volts and it passes a current of 55 mA, whereas the correct
figures are 250 volts and 48 mA. The probability is that there
is nothing wrong with this stage and that the unusual operating
conditions are due to the output of the mains equipment being
unusually high.
This is easily seen by running over the possible faults and
their symptoms. If the valve had lost its emission or its
filament voltage were low, the anode current would be below
normal ; this would mean a smaller load on the mains equip-
ment, and the anode voltage would consequently be normal to
extra high, depending on the H.T. regulation. If the bias
resistance had increased its value, the same thing would happen,
but the cathode voltage would also be high. Similarly, a
failure of the grid bias supply to reach the grid of the valve
would result in a heavy anode current with a low anode voltage.
If this defect were in the grid circuit of the valve, the cathode
volts would also be high, whereas if it were due to the bias
circuit they would be low.
WIRELESS SERVICING MANUAL
Checking the A.F. Amplifier
The matter is probably most clearly seen from Fig. 12,
which shows an A.F. amplifier embodying both resistance and
transformer coupling. The first step is to measure the anode
and cathode voltages of V3 from the earth line E, and the
difference betvveen them gives the true anode-cathode voltage ;
then check the anode current. If any abnormality cannot be
explained by the voltage of the H.T. line being high or low, the
trouble must lie in V3 or its immediately associated circuits.
Check the value of the bias resistance R8 with an ohmmeter,
or from the voltage and current readings, R (in ohms) = Volts
across R/Current through R in amperes. If the resistance is
too low, the resistance itself may not be at fault, but the by-pass
condenser Cy. If all be in order, short-circuit the A.F. trans-
former secondary. If this brings things back to normal an
open-circuit in the transformer secondary should be suspected
and the winding checked for continuity with an ohmmeter.
Should it be found continuous, the trouble is probably parasitic
oscillation, and grid and anode stopping resistances should be
inserted.
The Previous Stage
In the previous stage, the possible faults are greater in
number but just as easy to determine. Suppose the check on
the valve conditions reveals low anode voltage and current.
The trouble cannot then be due to Vz itself, for although low
heater voltage or a loss of emission would cause low anode
current, it would make the anode voltage abnormally high.
The voltage at the junction of the coupling and decoupling
resistances R5 and R4 should be checked, therefore. If it be
nearly the full line voltage, the trouble is an open circuit in
R5, but if it be below normal a leak in C4 or C5 should be
suspected.
In this connection, it should be remembered that if C4 is an
electrolytic condenser it is normal for it to pass a slight current,
up to about I mA. Should the anode voltage be practically
zero, but some voltage be present at the junction of R4 and R5,
a short-circuit in C5 is almost certain, but if the voltage at
both points is zero, then the short-circuit is in C4.
When the anode current is high, however, but the anode
voltage low, the trouble lies in the grid or cathode circuits.
The cathode circuit is exempt from suspicion if it be found
that the product of the bias resistance R7 (in ohms) and the
anode current (in amperes) equals the measured cathode voltage.
20
THE INTERPRETATION OF METER READINGS
Approximate agreement is good enough, for the measured
voltage may be somewhat below the true value owing to the
load of the voltmeter. If the measured voltage be very low,
check R7, C6, and the heater-cathode insulation of the valve.
It is useless to do this last with the heater cold. When the
cathode circuit is found to be correct, short-circuit the grid
leak R6. If the currents and voltages then become normal,
the trouble is a leaky condenser C2.
A very small leak in such a condenser can completely upset
the conditions of the following valve. Thus suppose that Vi
operates with an anode potential of 100 volts and that R6 is
0*25 megohm. Then a leak of 20 megohms in C2 will result
i n the grid of V2 being
100 X 0*25
1-24 volts positive with
20*25
respect to the earth line. If the cathode potential of V2 is
3 volts, the true grid voltage with respect to the cathode will
be not — 3 volts, but 1*24 — 3 = — 1*76 volts only. An
internal resistance lower than 20 megohms in C2 will naturally
result in a still lower effective grid bias on V2, and it may
become positive instead of negative.
Grid Circuit Conditions
The grid circuit conditions can always be checked by shorting
the grid directly to the source of bias, in this case by shorting R6.
If there is a change of anode current, then it is certain that
there is current flowing in the grid circuit ; either the valve
itself is passing grid current due to incorrect operating voltages
or to its being defective, or some other circuit is causing a
current to flow through the grid circuit.
It is readily seen that where resistance-coupling is used before
the output valve, any leak in the coupling condenser may
seriously damage the output valve and possibly the H.T.
rectifier as well, for all anode circuit components are of fairly
low resistance and a very heavy anode current may flow. For
this reason, the coupling condenser to the output valve is
sometimes of the mica-dielectric type. Damage to the valves
is unlikely through this fault in earlier stages, for the early
valves have high resistance anode circuits which limit the
possible anode current to a figure which the valve can easily
withstand.
The valves employed in R.F. and I.F. stages are usually of
the screened pentode type. With these the rules for circuit
testing are in no way different from those applicable to the
triode, save that it must be remembered that with all such
21
WIRELESS SERVICING MANUAL
valves operating under normal conditions the screen potential
exercises a much greater effect upon the anode current than
does the anode voltage. In the case of screened-tetrodes the
screen current is a very variable quantity, and one specimen may
pass two or three times the current of another equally good one.
Sometimes, too, the screen-current is reversed, or negative,
without the valve being in any way defective. Pentodes are
more consistent and the negative current effect is rarely, if
ever, found ; the screen current is higher, however, and in the
case of R.F. pentodes it is often about 2 mA. With an output
pentode, of course, the screen (really the space-charge grid)
current is usually 8-15 mA.
When checking voltages and currents in any part of the
receiver, but particularly when testing the pre-detector stages,
the volume control should be at maximum, unless it be a purely
A.F. control, and the set mistuned from any signal. Unless
this is done, many readings are bound to be inaccurate and will
lead one on a wild goose chase. If the set is tuned to a signal,
for instance, A.V.C. will bias back certain valves and reduce
their anode and screen currents and cause a rise in certain
voltages.
Resistance Testing
When voltage and current testing fails to indicate a fault,
or when it shows one to be present but fails to reveal its exact
nature, it may often be found by resistance testing. This is
merely an elaboration of continuity testing, using an ohmmeter
for the purpose. A break in the primary winding of an A.F.
transformer would certainly be shown up by the voltage and
current tests if the primary is directly fed, but not if it be
resistance-capacity fed in the modern way. It may be checked
instantly with the ohmmeter, however, simply by connecting
the meter leads to the primary terminals. Similarly with
the secondary, the ohmmeter will infallibly reveal a break in
the winding, whereas with other tests there is an element of
uncertainty.
The ohmmeter is of the greatest value, however, in the R.F.
circuits, for the operation of wavechange switches and the
continuity of coils may be readily checked. In a set covering
the medium- and long-wavebands only, waveband switching is
often arranged on the lines shown in Fig. 13 for one circuit,
and the switch is open for the long waveband. If the switch
be set for the long waves and the ohmmeter connected across
both coils at the points AC, it will show the total D.C. resistance
of both coils, usually about 10-40 ohms.
22
THE INTERPRETATION OF METER READINGS
Fig. 13 : A wavechange switch can
be checked by connecting an ohm-
meter at AC, to determine whether
the resistance changes on operating
the switch
Fig. 14: This diagram shows a typical
method of waveband switching for
an “ all-wave ’* set
Moving the switch to the medium waveband should short-
circuit the long wave coil, and the meter should indicate the
resistance of the medium wave coil, some 1-5 ohms. If the
meter reading does not change, obviously the switch is not
making contact. Even if a change of reading of the expected
order is obtained, however, it is not certain that the switch
contact is good.
This may be determined by connecting the meter directly
across the medium wave coil (AB) and noting the resistance.
The figure should be the same as with the switch closed and
the meter connected across both coils (AC). It cannot be
greater but may be less ; if it appears greater there is a bad
connection to the meter. The difference of the readings
represents the resistance of the switch contacts. A contact
resistance of 0‘i ohm may not materially impair the performance
of the receiver, though a resistance of i ohm will certainly do
so, but on the medium waveband only.
In multi-band sets the waveband switching usually takes the
form shown in Fig. 14. If the ohmmeter is connected across
AB, it will indicate different resistances on the medium and
long wavebands, and still lower resistance on the short waves.
The short-wave coils are of such low D.C. resistance, however
that the average ohmmeter will indicate zero. The test will
show up an open-circuit in a coil or switch contact, but will
not necessarily reveal a bad contact of sufficient magnitude
seriously to affect the short-wave performance. This is because
the tolerable resistance in a switch contact becomes smaller as
the coil resistance gets lower.
When checking primary and reaction windings the resistance
23
WIRELESS SERVICING MANUAL
may be found to be the same in both positions of the switch.
This does not always mean a fault, for in some sets these wind-
ings are the same on both wavebands. Again, in a few cases
such coils will be found to have a surprisingly high resistance,
perhaps 250 ohms. This is because primaries and reaction
coils are sometimes wound with resistance wire in order to
prevent parasitic oscillation and, in the case of an oscillator, to
keep the output at a fairly even level over the waveband.
In order to avoid false readings
and the possibility of damage to the
I '*' meter, the set should always be
switched off, or if a battery set,
^ I disconnected from its batteries , when
using an ohmmeter. Care should
also be taken to see that the circuit
under test is really the only one
^2 connected to the meter, for there
are many cases where an alternative
. path may exist. The meter cannot
tell the difference between different
Si o"iy ‘i>ow.
always be disconnected. If left resistance between the two points to
in circuit misleading results may which it is connected,
be obtained through the presence case of the bv-paSS
of resistances , • • j
condenser in the screen-grid
circuit, as shown in Fig. 15, for instance. If an internal leak
is suspected, it is of little use connecting the ohmmeter across
it while it is still connected to the voltage divider, for the meter
will in any case read the resistance of Rz. It should be dis-
connected at one end and checked alone. If the test is for a
short-circuit only, however, then disconnection is unnecessary.
The meter will give a reading, it is true, even if the condenser
is in order, but it will show the value of Rz — 5,000-20,000
ohms — whereas with a short-circuited condenser it will read
0-100 ohms.
When searching for open circuits, the ohmmeter is just as
useful, for when connected to the circuit at fault it will continue
to read infinity instead of showing the correct resistance. It is
most useful when searching for a disconnection in the wiring,
for there are many cases where a joint may be passed by a
visual inspection as sound, but a resistance test immediately
reveals it to be defective.
24
CHAPTER 4
VALVES
W HEN a fault develops in a receiver, it is the valves
which are usually first suspected because they are not
only subject to failure like any other component but
they most definitely wear out with use. Unfortunately, they
are much more difficult to test properly than are most com-
ponents. Unless proper testing equipment is available, the
best course is undoubtedly to substitute a new valve in the
receiver.
As there are over 600 current types, to say nothing of the
American valves used in imported sets or the obsolete types
which are still used in many old receivers, this method is
obviously out of the question in general servicing. Where only
a moderate number of types of receiver is handled, of course,
it is quite possible to keep a set of suitable valves in stock and
this is to be strongly recommended.
Proper testing gear which can be relied upon to detect any
fault in a valve is complicated and expensive. As the majority
of faults can with a little skill be detected while the valve is in
the set, this course will appeal to the majority. Short-circuits
between the electrodes, or a broken heater or filament can, of
course, be readily detected by connecting an ohmmeter to the
various valve pins.
Defective heater-cathode insulation cannot often be detected
in this way, however, for it usually appears only when the
cathode has attained its normal working temperature.
Sometimes, too, it is intermittent and may not appear until the
set has been in operation for an hour or so, and then only for
a few minutes. In most circuits, the bias resistance is connected
between cathode and chassis, and the centre-tap on the filament
winding of the transformer is joined to the chassis. Poor
heater-cathode insulation consequently partially or completely
short-circuits the bias resistance and the fault is most easily
detected with a voltmeter between cathode and chassis, for
when the breakdown occurs the cathode voltage becomes
much below its normal figure. If the fault is intermittent it is
naturally much more difficult to trace, and the only course is
to leave the voltmeter in circuit until it occurs.
Microphonic and noisy valves can generally be detected by
tapping them, but with certain kinds of noise, notably hiss, the
only certain test is replacement with a good specimen. A
B 25
WIRELESS SERVICING MANUAL
valve which introduces mains hum does not normally need
special tests for it will be located during the process of carrying
out the normal hum tests described in a later chapter.
Emission Falling Off
The end of the normal life of a valve is generally set by a
falling off in the emission. With a battery set, this may often
be detected through the anode current being abnormally low
for the voltages applied to the valve. With a mains set, how-
ever, this test is of little use, except when the valve is in the
last stages of decrepitude, for the automatic grid bias circuit
tends to keep the anode current up, and the current may not
be much below normal when the valve has deteriorated
considerably.
The best test is undoubtedly the measurement of mutual
conductance. This is not as difficult as it sounds, and rough
measurements can be made quite easily. The mutual conduct-
ance is actually a change of anode current divided by the change
of grid voltage necessary to produce it. If arrangements be
made to operate the valve under suitable conditions and at a
certain grid bias, therefore, the difference between the anode
currents obtained at this bias and when it is changed by i volt
gives the mutual conductance directly in mA/V.
The circuit of a suitable valve tester is shown in Fig. i6 and
with triodes the standard voltages for tests are loo volts H.T.
and zero grid bias. The slider of the potentiometer, which
should be of fairly low
resistance, say 500 ohms,
should be set so that it
is I volt negative with
respect to negative H.T.
The switch S should be
set to the upper position
for zero grid bias and
the anode current noted ;
the switch is then thrown
to the lower position
and the new value of
anode current noted.
The difference between
the two readings gives
the mutual conductance.
Thus, if the current
with zero grid volts is
Fij. 16 : A simple valve-tester can be built with
this circuit. If the switch changes the grid bias
by I volt the change of anode current (mA) gives
the mutual conducunce (mA/V) directly
26
VALVES
lo mA and with i volt bias it is 8*3 mA, the mutual conductance
is 10-8*3 =1*7 mA/V. The arrangement may, of course, be
used also with screen-grid, R.F. pentode, and output pentode
valves, but it is necessary to provide a tapping on the H.T.
battery of suitable voltage for the screen.
If many valves have to be checked it is a good plan to wire
up a number of valve-holders permanently, one for each
different class of valve, together with suitable batteries, meter,
and switching. Any valve can then be given a rough test for
mutual conductance at a moment’s notice.
A still simpler method of measuring mutual conductance is
possible when an A.C. supply is available. The D.C. meter of
Fig. 16 is replaced by an A.C. milliammeter connected to the
anode circuit through a suitable transformer, and a known A.C.
voltage is applied to the grid of the valve by means of a trans-
former. The meter then reads the A.C. anode current produced
by a known A.C. grid voltage and it can be calibrated directly
in mutual conductance. If the tester is kept permanently
wired up to suitable batteries, therefore, it is only necessary to
plug in a valve and read off its mutual conductance from the
meter, no switching or calculation being necessary. The
27
WIRELESS SERVICING MANUAL
method is not, of course, highly accurate, but it is convenient
and quite good enough to enable good and bad specimens to
be distinguished with certainty.
For accurate results, of course, a proper bridge is necessary.
This need not be complicated, however, for good results can be
obtained with fairly simple equipment such as that described
in Appendix IV. Apparatus of this nature enables accurate
measurements of mutual conductance to be made over the wide
range of io*o mA/V to o*oi mA/V.
‘Avo” Valve Characteristic Meter
28
CHAPTER 5
TRACING MAINS HUM
A lthough mains hum is a particularly annoying defect
in a receiver, it is one which is usually easy to cure.
The number of points at which hum may enter a receiver
is limited, and the search for the cause of the trouble, whether
the receiver be new or old, resolves itself into a systematic
checking of the various circuits. Fundamentally the problems
encountered are the same in A.C., D.C., and A.C./D.C. sets,
but the two last sometimes present special difficulties not met
with in purely A.C, models.
The simpler case of an A.C. set will be considered first,
therefore, for most of the treatment is applicable to other
types. The method of tracing hum is the logical one of stage
by stage testing, starting at the loud-speaker and working
backwards to the aerial ; if the process be carried out correctly
this system infallibly reveals the cause of the trouble, and then
it is usually easy to devise a remedy. The procedure is best
illustrated by an example and the circuit of Fig. 17 forms a
good basis for discussion. This circuit is not one which is
now commonly employed, although it has been widely used in
the past ; it is, however, particularly suited to our present
purposes, since most of the different forms of mains hum can
occur with it.
The first step in testing is to absolve the loud-speaker from
blame, for it is obvious that 'if the hum is occurring in this it
is a waste of time to investigate the receiver proper. Hum in
the loud-speaker occurs when the current energising the field
winding has an appreciable ripple on it and a hum-bucking
coil is not fitted.
In order to determine whether or not such hum is present,
disconnect the output transformer primary and join together
the two leads which are normally connected to it, in order that
H.T. may be applied to the output valve and the load on the
mains transformer maintained at its normal figure. Then
connect a resistance across the transformer primary having a
value approximately equal to the internal A.C. resistance of
the output valve, so that the transformer is normally loaded.
The A.C. resistance can be found from the valve-maker’s
catalogue or from the Wireless World Valve Data Supplement.
Then switch on the set and listen for hum. If it be present,
it is occurring in the field or is being picked up by the output
29
WIRELESS SERVICING MANUAL
30
Fig. 17 : Thii circuit diagram, although not one which is widely used nowadays, is particularly well adapted for illustrating the
methods of tracing hum
TRACING MAINS HUM
transformer. The second is very unlikely, but the first is
fairly common.
The cheapest remedy is to send the speaker back to the
makers for the addition of a hum-bucking coil ; this usually
removes 90 per cent, of the hum, but it often leaves a trace of
fairly high-pitched hum, which is often not enough to be
troublesome. The alternative remedy is to smooth the field
current ; if sufficient smoothing is used this is a certain and
complete cure.
The Speaker Field Supply
With the circuit of Fig. 17, hum of this nature should not
occur, for the field current is smoothed by CH2 and C12, which
also provide the smoothing for the output stage and initial
smoothing for the early stages. If hum does occur with an
arrangement of this nature, it probably means that C12 or CH2
is defective, but the absence of speaker-field hum is not proof
that these components are in order.
In practice, this type of hum occurs chiefly when the field
is separately energized, but it may occur in cheap sets where
the field winding forms the only smoothing choke in the set
and the speaker is not well hum-bucked. In such cases the
best remedy is to insert a choke in series with the field winding,
on the rectifier side of the winding, and to connect a condenser
to negative H.T. from their junction.
This is shown in Fig. 18, in which Ci and C3 are the con-
densers normally fitted, and CH and C2 are the new components.
Fig. 18 : Hum which comes from the speaker field supply can be stopped by
connecting a choke and condenser before the field
31
WIRELESS SERVICING MANUAL
Fig. 19 : With a separately energized speaker smoothing may be needed to
prevent hum. When the field is of low voltage (a) the condensers should be
200 /iF or more, and the choke of only I or 2 ohms resistance, but with a high-
resistance field (b and c) 8 condensers are suitable with an ordinary choke
The choke need have an inductance of no more than 8 H. or
lo H. at the maximum current taken through the field, but it
must be of low D.C. resistance if it is not to drop the H.T.
voltage considerably. For convenience the condenser C2 will
in most cases take the form of an 8 ftp electrolytic.
Where the field is separately energized, the commonest
arrangements are shown in Fig. 19. A low- voltage heavy-
current field (a) is often very difficult to treat, for the resistance
of the smoothing choke can rarely be allowed to exceed i or
2 ohms : the inductance, therefore, can only be a fraction of a
henry and capacity must chiefly be relied upon for smoothing.
Fortunately, electrolytic condensers of 200 fiF or so can be
obtained quite cheaply.
32
TRACING MAINS HUM
Very largely on account of the difficulty of eliminating hum,
low- voltage fields are now little used, and the usual arrangement
is that of Fig. 19 (b) with a field having a resistance of 2,000-6,000
ohms. Here smoothing is easy and a 10 H. choke and 8 /^F
condenser should readily remove all traces of hum. If they do
not, try the choke and condenser on the other side of the field,
as in (c ) , for as no mains transformer is used the results may be
affected by the earthing system of the mains.
When hum from the speaker field has been proved to be
absent, or remedied if present, the connections of the output
transformer should be replaced to normal. If hum is now
present, short-circuit the input circuit of the output valve —
the A.F. transformer secondary in Fig. 17 — or the grid leak in
the case of a resistance amplifier. If the hum continues the
defect lies in the circuits still operative, whereas if it ceases it
is obviously coming from an earlier stage.
The Output Stage
Assuming that the hum continues, it can only be due to the
output valve, or its H.T., bias, or filament supplies : in the
case of a pentode, there is also the space-charge grid H.T.
supply. The anode supply can be tested by removing the
output valve and connecting a resistance between negative
H.T. and the anode socket of the valve-holder ; the value of
the resistance should be such that the normal current flows
through the output transformer. With a PX4 valve, for
instance, the anode voltage is 250 volts and the current 48 mA ;
its resistance to D.C. is thus 250/0*048 = 5,200 ohms. The
resistance, of course,
should be of at least 12
watts rating, and its exact
value is unimportant. In
this case, 5,000 ohms
would be quite near
enough.
If the hum continues
with this resistance in
place of the valve, it can
only be due to inadequate
smoothing of the H.T.
supply for this stage, and
the choke and condensers
should be checked . I n ex-
treme cases an additional
Fig. 20 : Hum from the grid bias source may
often be prevented by grid-circuit decoupiing
33
WIRELESS SERVICING MANUAL
stage of smoothing may be needed, but these should be very rare.
When silence has been secured, replace the valve, and if there
is again hum it can only be due to the bias or filament supplies.
If the bias resistance by-pass condenser, C14 of Fig. 17, is
over 25 mF, there should be no hum from this source and it
is an easy matter to check it by temporarily connecting another
in parallel.
If the condenser is small, however, it may be necessary to
replace it by a large capacity or to decouple the grid circuit.
The latter can only be done with transformer coupling and the
connections are shown in Fig. 20, where R2 is the bias resistance
and Ri the decoupling resistance of some o*i megohm. The
condenser C should be about i mF ; it must not be of the
electrolytic type, since the leakage would reduce the effective
grid bias with ruinous effects on the life of the valve.
Hum from the filament supply is likely only if the tapping
on the mains transformer secondary is much out of centre or
if the valve used has a low-current filament. In either case,
the best results will be secured by using a potentiometer across
the filament, instead of the centre-tap ,as shown in Fig. 21,
where Ri is the bias resistance and Ci is the by-pass condenser.
The potentiometer P should have a value of some 15-30 ohms
and the slider should be adjusted for minimum hum.
Miniature potentiometers for this purpose are obtainable
commercially from various firms, including Claude Lyons Ltd.
In general, however, such potentiometers are needed only when
the defect lies in the mains transformer tapping being off-centre.
With modern heavy-
current valves, they are
usually unnecessary,
and almost invariably so
with indirectly heated
valves. Filament hum
is usually serious only
with the old 0*25-
ampere type valves, and
if the use of a potentio-
meter does not cure it
the only remedy is to
change the valve for one
consuming a heavier
Fig. 21 : The use of a potentiometer P across the filament current,
valve filament will tend to reduce hum from the With a oentode OUt-
LT supply when the mains transformer secondary P, ,
is not accurately centre-upped pUt valve, the
34
TRACING MAINS HUM
space-charge grid repre-
sents an additional
possible source of hum
and one which is, in fact,
the most probable of all.
The space-charge grid
H.T. supply requires
more smoothing than
that of the anode, for
any ripple on the supply
is amplified to some
degree by the valve. It
is the common practice
to feed this electrode
from the same point
in the smoothing equip-
ment as the anode, and
when this is satisfactory
it really means that the
anode supply is
smoothed rather more
than is strictly necessary.
When the anode
supply is only smoothed just enough for hum to be avoided
in this circuit, more smoothing is needed for the space-
charge grid. In some cases, such additional smoothing can
be secured by feeding the grid from a later point in the
smoothing equipment ; in others a simple filter is needed. Thus
in Fig. 22 the insertion of a resistance R in the space-
charge grid lead, with a by-pass condenser C, will usually
effect a cure. The resistance R should be no larger than
necessary, for it lowers the grid potential and so reduces the
power output. It should be from 5,000 to 10,000 ohms and
C can well be 8/iF, although smaller capacities will sometimes
suffice. Where the voltage drop in a resistance cannot be
permitted, R must be replaced by a choke. This is less
expensive than it sounds, for as the current is only 8-10 mA
quite ar high inductance can be obtained cheaply. Whatever
type of valve be used, it should always be remembered that in
a push-pull output stage, serious mismatching of the valves
may be a cause of hum. Since such a stage is not normally
responsive to a ripple on the H.T. supply, less smoothing is
often used than would be necessary with an ordinary output
stage.
Fig. 22 : The space-charge grid of a pentode
sometimes needs additional smoothing, which
can be effected b/ a condenser C and resistance
R. This resistance must not be of too high
value otherwise there will be an excessive
voltage drop across it
35
WIRELESS SERVICING MANUAL
The A.F. Transformer
When satisfied with the performance of the output stage the
short-circuit on the input should be removed, and if hum now
reappears the preceding stage must be tested. If transformer
coupling is used, as in Fig. 17, the hum may be picked up by
this component. This is easily tested, and the procedure is to
disconnect the primary and to connect across it a resistance
roughly equal to the valve resistance (V3) in order to simulate
working conditions.
If hum is now present which disappears on short-circuiting
the secondary, it is quite certain that it is picked up by the
transformer — usually from the mains transformer but sometimes
from smoothing chokes. One remedy, of course, is to separate
the components more widely, but this is often impracticable.
Screening is of little or no value, and the only course is to
orient the A.F. transformer to the position of minimum hum.
The transformer should be unscrewed from the chassis and
flex leads fitted for the secondary connections. It can then be
readily moved about in the set and the position of minimum
hum determined ; this position is very critical if the mains and
A.F. transformers are close together, but may not be if they are
more than a foot or so apart. Quite probably, the transformer
will have to be tilted slightly, and as in some cases several
minima can be found, the one chosen will naturally be that
which gives the shortest leads or the easiest mounting.
Having mounted the transformer in such a position that the
hum is negligible, the connections may be replaced to normal,
and the operating conditions of the valve investigated. When
the set is switched on again, it may be found that hum is again
present but that this dies away as the valves warm up. This
is due to hum pick-up in the transformer. Even when set for
minimum hum, the pick-up is rarely zero and depends
largely upon the degree of damping imposed upon it by
the valve.
The lower the resistance of the valve feeding the transformer,
the less likely is there to be hum, and it is for this reason that
the connection of a resistance across the primary is recommended
when orienting the transformer. If no resistance be used, the
minimum may be ill-defined and is unlikely to be silent. Under
working conditions, the transformer is damped by the valve
resistance and correct operation is secured. When the set is
first switched on, however, the resistance of V3 (Fig. 17) is
infinite, for the valve cathode takes 15 to 30 seconds to acquire
its working temperature.
36
TRACING MAINS HUM
The directly heated
output valve is operative
almost at once, how-
ever, and hum is heard
because the transformer
is not damped by V3.
As this latter valve
warms * up, the hum
decreases, and ceases to
be audible when it is
working normally.
This effect cannot be
avoided without sepa-
rating the transformers
very greatly ; but it
may be reduced by
using parallel-feed con-
nections for the trans-
former. These are
shown in Fig. 23.
Under normal condi-
tions, the damping on
the transformer primary is due to Ri in parallel with the valve
resistance, and when the valve heater is cold, to Ri alone.
Some change of damping is experienced as the valve warms up,
of course, but it is much less than with a directly-fed
transformer.
Fig. 23 : The connections for a resistance-
capacity fed A.F. transformer
The A.F. Stage
Reverting to the question of hum during normal operation,
when pick-up in the transformer has been satisfactorily
eliminated, the next step is to test the circuits of the valve. Its
input circuit should be short-circuited (R9 of Fig. 17), and hum
can then be due only to the valve, or its anode or bias supplies.
It is usually possible to tell whether the valve is at fault or not
by the character of the hum. If it is of fairly low pitch and
perfectly steady and unchanging, the valve is unlikely to be at
fault, but if the hum wanders in pitch or intensity, and particu-
larly if it is of comparatively high pitch, then the trouble is
almost certainly a faulty valve.
Assuming the valve to be in order, hum indicates a lack of
smoothing in the H.T. supply, or inadequate by-passing of the
bias resistance. The latter can be checked by trying a 50^ F
37
WIRELESS SERVICING MANUAL
condenser across it (C9, Fig. 17). If the H.T. smoothing were
initially in order, the hum can be due only to an open circuit
•in, or bad connection to, the decoupling condenser for this
stage (Cio) or one of the smoothing condensers (Cii, C12) or
to short-circuited turns on one of the smoothing chokes. If
the receiver has never been satisfactory from the point of view
of hum, the same possibilities apply, but the trouble may
really be due to insufficient smoothing equipment ’and an
additional stage may be needed.
In this connection, it may be remarked that decoupling
circuits give a very appreciable amount of smoothing, and hum
may sometimes be remedied merely by increasing the capacity
of a decoupling condenser.
When silent operation has been secured, the short-circuit on
the input should be removed, and any hum now found must be
coming from an earlier stage. If this is transformer coupled,
the procedure is exactly the same as before ; if it is resistance
coupled the procedure is as follows, and it should be noted that
this applies to any resistance-coupled stage and not merely to
the precise arrangement of Fig. 17.
The first step is to remove the preceding valve (V2) and to
short-circuit the R.F, choke (CHi), if any. If hum be present,
disconnect the coupling condenser (C8) from the coupling
resistance (R7). If this removes the hum it is undoubtedly due
to the H.T. supply to V2 and a defect in the smoothing or
decoupling circuit is indicated. The condensers should be
checked, but if there is no defective component, an increase in
smoothing or decoupling capacity is needed ; an increase in the
decoupling condenser (C7) will usually be satisfactory.
The R.C. Coupled Stage
Should the hum be present with the coupling condenser (C8)
disconnected, however, it is being picked-up on the grid and
grid circuit components of V3 from the electric hum-field
rather than the magnetic. A reduction in the value of R9 will
usually help, but this will reduce the bass response unless
accompanied by a proportionate increase in the capacity of the
coupling condenser C8. For a given bass response, the actual
values of coupling condenser and grid leak are unimportant,
and it is only necessary that their product should be constant.
A reduction in R9^ however, will reduce the amplification
somewhat, so one cannot go far in this direction. The true
remedy for hum pick-up of this kind is screening. A metallized
valve should be used, the grid leads made as short as possible,
38
TRACING MAINS HUM
and run in screened
sleeving if they are
more than an inch or
so long.
When hum has been
eliminated the preceding
valve can be replaced
and its input circuit
short-circuited. Any
hum which now occurs
can be due only to the
valve or its bias supply,
if any, and the procedure
for removing this has
already been given.
When silence has once
again been obtained,
remove the short-circuit
from the R.F. choke, if
there is one ; if this introduces hum it is being picked up by
the choke. One remedy is to orient the choke to the position
of minimum hum, just as in the case of an A.F. transformer,
but it need not, of course, be disconnected. The alternative
is to replace it by an astatically wound choke, such as a
binocular choke.
The next step is to remove the short-circuit from the input
and treat the preceding stage in the same manner, if it is
another A.F. stage. If this valve is a grid detector, however,
and transformer coupling is used between it and the preceding
valve as in Fig. 24, any hum now introduced can only
be picked-up on the detector grid from the electric field, for which
the remedies have already been given, or pick-up by the tuning
coil from the mains transformer. This last is only likely if the
coil is of very large diameter, or is an I.F. transformer having
high inductance coils : even then, it is only likely if the coil
is mounted right alongside the mains transformer.
When tuned anode coupling is used, however, as in Fig. 17,
a ripple on the H.T. supply to the R.F. valve may cause serious
hum, for an appreciable proportion of it will be passed straight
to the detector grid in spite of the small size of the coupling
condenser C4. The remedy, of course, is to increase the
smoothing of the H.T. supply to this valve, or to look for a
defect in the smoothing system.
It should be noted that this type of hum is quite probable,
39
WIRELESS SERVICING MANUAL
for the anode voltage of the R.F. valve is often less wrell smoothed
than that of the detector, since the supplies are taken from the
same point on the H.T. supply system, and less decoupling is
used in the R.F. stage. This particular variety of hum does
not occur with the tuned grid circuit any more than it does
with transformer coupling, but the ripple may cause trouble in
other ways.
The testing procedure outlined in the foregoing will, if
faithfully followed, reveal the cause of any hum of the type
which is continually present whether the receiver be tuned to
a signal or not. It is, however, rarely necessary to go through
the whole procedure, and indeed it would be unutterably tedious
to do so. The full process is necessary only when every type
of hum is simultaneously present, and it would be a very
exceptional set in which this occurred.
It is not uncommon for a set to have several sources of hum,
but their location is not usually a difficult or a lengthy business.
Hum is, of course, more likely to occur in a newly-constructed
receiver than as the result of a breakdown, except in the matter
of valves. It may often be necessary to remove it from old
sets, however, for many of these gave appreciable hum which,
though tolerable at the time, would not be so to-day.
Localising the Trouble
The short-cuts to testing consist of a few rapid tests which
localise the source of the trouble to one portion of the receiver.
If a receiver is arranged to operate with a pick-up, for instance,
the first step is to short the pick-up terminals and turn the
control switch to gramophone. If the hum ceases, one knows
at once that the circuits following the pick-up switch are in
order and that it is a waste of time to investigate them ; it
should be pointed out, however, that this test does not necessarily
absolve the grid circuit of the valve to which the pick-up is
connected. Should the hum continue unabated, it is obviously
occurring in a later circuit.
As an example of the method of rapid testing, suppose that
a receiver of the type of Fig. 17 is giving trouble from hum.
The first thing is to short-circuit the A.F. transformer secondary.
If there is hum, investigate the speaker field, and if that fails,
the output stage itself. If the hum ceases, however, one knows
at once that the output stage and field supply are in order, and
the next step is to remove the short-circuit from the transformer
secondary and apply it to the input of V3. Suppose this
reduces the hum, but does not completely remove it. It is
40
TRACING MAINS HUM
clear that some of the hum is occurring in this stage and some
at an earlier point. The A.F. transformer should be tested for
hum pick-up and if this fails, the H.T. supply and valve.
When this particular source of hum has been removed, and not
before, the short on the input can be removed and transferred
to the preceding stage. The remaining hum is in the extra
stage now in circuit and a few tests on the lines described will
speedily reveal its source.
Gramophone Hum
In the case of gramophone equipment, there are additional
sources of hum ; the pick-up itself and the leads to it. Little
extra testing is needed, however, for the pick-up leads should
invariably be screened, and if the pick-up itself appears respon-
sible for the hum the real cause is usually a faulty or badly-
designed motor. Care should, of course, be taken to see that
the motor frame is earthed and that the insulation of the motor
windings from the frame is satisfactory. A dirty commutator
will cause serious noise which can hardly be described as hum,
but excessive vibration may lead to hum as soon as the pick-up
is placed on the record.
Before leaving the question of direct mains hum, it may be
as well to point out one or two ways of increasing the smoothing
with little additional apparatus. In an A.C. set, the hum is of
definite frequencies, and where one frequency predominates it
is possible to reduce it greatly by using a tuned smoothing
circuit.
Where full-wave rectification is used, the chief hum frequency
is twice the mains frequency, or loo c/s in the case of 50 c/s
mains. If one of the smoothing chokes is shunted by a con-
denser of the correct value it will resonate at 100 c/s and very
greatly reduce the hum. When the inductance of the choke is
known the condenser capacity can readily be calculated ; it is
given by C = i/c^^L where C is the capacity in farads, L is
the inductance in henrys, and = 6*28 X frequency. Thus,
to tune a 10 H. choke to 100 c/s, the capacity must be 0*254
Tuned Smoothing System
There is one grave disadvantage about this arrangement
which renders it of only occasional use ; — although it greatly
increases the smoothing at the resonant frequency, it reduces
it at other frequencies. The chief hum frequency is 100 c/s,
but there are also frequencies of 150 c/s, 200 c/s, 250 c/s, and
41
WIRELESS SERVICING MANUAL
so on, and although they may be weaker, both the loudspeaker
and the ear respond to them more readily. Thus, while tuning
a choke may remove lOO c/s hum, it may introduce hum of
higher frequency, so that one is no better off. There are,
however, occasions where the system is of benefit.
When half-wave rectification is used, as in most Universal
receivers, the chief hum frequency is at the mains frequency,
usually 50 c/s, but there are multiples of this frequency. It is
definitely dangerous to use tuned smoothing chokes in such
sets, even if they prove advantageous on A.C. The hum
frequencies with D.C. mains are usually quite high, and a
tuned smoothing system which is suitable for A.C. will offer
little barrier to the ripple on a D.C. supply.
Tracing hum in D.C. and A.C. /D.C. sets is much the same
as in a purely A.C. set. It is rendered more difficult by the
fact that one cannot remove valves while testing since their
heaters are wired in series, but to compensate for this there is
no mains transformer from which other components can pick
up hum. The chief difficulties are likely to be found in cases
where the positive side of the mams is earthed, and in con-
tradistinction to the usual case with A.C., where a good earth
is an advantage and sometimes an essential, better results are
often secured without any earth connection at all.
One of the main troubles with D.C. supplies is that the
same remedies do not apply in different cases. Hum troubles
in the set itself, of course, respond to the usual tests just as in
the case of an A.C. set, but one can never be certain that the
manner in which the set is used will not cause trouble. It is
unfortunately necessary to treat each case on its merits and one
can never guarantee that a D.C. set, however well it may be
designed, will be hum-free on all supplies. It is largely for
this reason that set designers have such an aversion to D.C.
receivers.
In connection with the use of a gramophone pick-up with
A.C. /D.C. and D.C. sets, it may be remarked that in certain
cases it is disadvantageous to employ earthed screening for
pick-up and other connections. Indeed, such screening may
lead to more hum than unscreened leads. Screening is, of
course, necessary for the avoidance of hum, but experience
indicates that it must not be earthed, but maintained at the
same potential as negative H.T.
It cannot be connected directly to negative H.T., however,
for screened leads are usually in accessible positions and there
would be a serious risk of shock or fire if they were in direct
42
TRACING MAINS HUM
connection with the supply mains. The best course is con-
sequently to connect all such screening to negative H.T.
through an isolating condenser which should not exceed o*oi /xF
for A.C. supplies and which should be rated for working at
300 volts.
Modulation Hum
Some of the most difficult forms of hum to trace appear only
when a signal is tuned in, and in the absence of a carrier the
receiver may be quite silent. Such hum arises because for
some reason the hum voltages modulate the carrier of the
station to which the receiver is tuned, and it is consequently
generally known as modulation hum. A ripple on the H.T.
supply to an R.F. or I.F. valve, for instance, may cause modula-
tion hum whatever the nature of the intervalve coupling. In
the case of tuned anode coupling, a ripple on the supply to the
last R.F. valve will cause direct hum, as already pointed out,
but it will not cause such hum with transformer or tuned grid
coupling. The hum may be impressed upon the carrier of a
signal, however, for the anode voltage of the valve will fluctuate
at the hum frequency, and unless the valve be perfectly linear
its amplification will vary in sympathy. Consequently, the
carrier voltage at the detector will be varying in amplitude at
the hum frequency ; in other words, the carrier will be modu-
lated by the hum.
Testing for Modulation Hum
The liability of a receiver to such hum is most readily tested
by connecting the output of an oscillator to the input of the
last R.F. or I.F. stage and tuning it to the appropriate frequency.
The oscillator should be unmodulated, otherwise any hum will
be obscured. The oscillator output should be increased until
hum appears.
If the input is judged to be larger than any signal input
likely to be applied to this valve under normal operating con-
ditions, nothing need be done, for the appearance of the hum
is due chiefly to the valve being overloaded and the ripple on
the H.T. supply is small. If the hum is prominent with a
normal input, however, the ripple is excessive and more smooth-
ing of the anode or screen supplies is indicated ; the stage is
incorrectly designed and cannot handle the input without
overloading, so that the anode and screen voltages should be
raised ; or the stage is overloaded through the deterioration of
the valve .
43
WIRELESS SERVICING MANUAL
By transferring the test oscillator stage by stage towards the
aerial, each stage can be tested individually for modulation
hum. In a receiver which has previously functioned satis-
factorily, the only likely causes are faulty valves, both in emission
and in heater-cathode insulation, and defective smoothing or
decoupling equipment.
The commonest cause of modulation hum, however, cannot
be found in this way. The hum appears usually only when
the receiver is tuned to the local station, and while such hum
may be due to overloading of an R.F. or I.F. valve, it is more
commonly caused by R.F. currents in the supply mains. Such
hum does not appear in A.C. sets which have a screened primary
mains transformer, provided that other coupling between the
mains and the receiver circuits is absent. The first step should
be to make sure that the screen is properly earthed ; if it is,
and it is found that reversing the mains plug in its socket
alters the degree of hum, the trouble is almost certainly due to
the wiring to the on-off switch. Screening the wiring to this
switch may effect a cure, but it is better to fit a new switch
well removed from the receiver circuits. The usual switch
attached to the volume control is in a particularly dangerous
position from the point of view of mains hum, for the mains
leads must necessarily pass through much of the receiver wiring.
In cases where the primary of the mains transformers is not
screened, it is obviously impracticable to attempt to introduce
screening and an alternative remedy must be found. A com-
plete cure will usually result from the connection of condensers
between the mains and
earth, as shown in Fig.
25 ; the value used for
these condensers C is
not critical, and o*ooi mF
is usually quite large
enough. The con-
densers should, however,
be rated for 250 volts
A.C. working.
Moving the
Receiver
When obstinate cases
of modulation hum
are found, it is always
44
TRACING MAINS HUM
Fig. 26 : Condensers across the high-voltage v/inding of the mains transformer are
helpful when the rectifier is responsible for hum
worth while to try operating the receiver somewhere
else. The house wiring is carried out in ceilings, floors and
walls, and it may occasionally happen that the receiver is used
in such a position that it is almost surrounded by concealed
house wiring and so may be in quite a strong hum field.
Similarly, the aerial lead-in may pass through such a field and
so lead to difficulty.
It is possible for the rectifier to give rise to modulation hum
in certain circumstances, although it is not very common.
The connection of condensers C as in Fig. 26 will usually cure
trouble of this nature and o-i fxF is the usual value. The
condensers must, however, be of very high voltage rating ; in
the case of Fig. 26 (t?) for one-half the total A. C. voltage of the
H.T. secondary, that is, 350 volts A.C. working for a 350-0-350
volts winding, or nearly 500 volts peak. With the arrange-
ment (6), however, even higher voltages are encountered, for the
voltage on the condenser is the sum of one-half the total A.C.
voltage and the rectified D.C. voltage. As the latter may be
nearly equal to the peak A.C. voltage when the set is first
switched on, the condensers should for safety be rated for
operation at the total A.C. voltage of the high voltage winding,
that is, 700 volts R.M.S. or 1,000 volts peak, for a 350-0-350
volts transformer.
With D.C. and A.C. /D.C. receivers, modulation hum can
again be caused by R.F. currents in the supply mains and its
45
WIRELESS SERVICING MANUAL
Fig. 27 ; In D.C. and Universal sets, a simple mains filter may be needed to
prevent modulation hum. The condenser C| is needed only when it is necessary
to use large capacities for C, and it should be of some 0.001 fiF capacity
elimination is much more difficult, for there is no mains trans-
former to give any isolation. The remedy, of course, is to
insert a filter in the mains leads, and in some cases condensers
to earth as in Fig. 27 (a) suffice ; for D.C. only the condensers C
should be about 0*5/1 /iF, but for A.C. it is inadvisable to
make them larger than o*i fiF. If large condensers are
necessary, then it is wise to insert Ci of o*ooi ^F to o*i fiF in
the earthing lead. It will often be necessary to include R.F.
chokes in the mains leads as well as these condensers. They
must be wound with heavy-gauge wire. No. 22 or heavier ;
suitable chokes for the medium waveband can be made by
winding 60 to 100 turns of No. 22 D.C.C. as a single layer on
a 2-in. diameter former. Fortunately, coupling between the
two chokes seems to have no adverse effect, but they should,
of course, be kept as far as possible from the receiver circuits.
It is rare for the heater supply to introduce hum in purely
A.C. sets, except on the short wavelengths where modulation
hum may be experienced. In general, such hum can be cured
by connecting by-pass condensers to earth from the heater of
the offending valve. The trouble usually arises in the oscillator
circuit, and 0*01 /uF mica or ceramic dielectric condensers are
to be recommended for the purpose. In A.C. /D.C. sets, the
order in which the heaters are wired is of importance, but there
is rarely any difficulty if the detector be placed at the earthy end
of the chain.
<6
CHAPTER 6
MOTOR-BOATING
T he name of “ motor-boating is usually given to that
form of instability which manifests itself by producing a
noise which is not unlike that of a petrol engine. It is
self-oscillation of very low frequency ; the frequency is rarely
higher than 25 c/s and is sometimes lower than i c/s ; it usually
lies about 2-5 c/s. Such extremely low frequencies are not, of
course, directly audible, but the amplitude of oscillation is so
great that the loudspeaker, and indeed the valves, are badly
overloaded, with the result that higher frequencies within the
audible range are present.
It is better to think of motor-boating as producing periodic
pulses of current or voltage than as generating an oscillation in
the common sense of the word. The amplitude reached is
often sufficient for the movement of the loudspeaker cone to
be readily visible, and if prolonged, it may damage the cone
suspension.
There are three distinct forms of motor-boating, and it is
necessary to distinguish them carefully if a remedy is to be
found easily. The three forms are (a) pure A.F. feed-back,
(b) R.F. instability, (c) modulation by A.F. feed-back. The
first is the most common and the easiest to deal with. It
invariably occurs through common impedance coupling, and
the impedance is usually that of the H.T. supply, although it
may occasionally occur from the source of grid bias. It is a
fault which in normal circumstances only occurs in sets deriving
their H.T. supply from a source other than batteries.
It is easy to see that the impedance of the H.T. supply is
included in the anode circuits of all valves. Referring to
Fig. 28, the impedance of the H.T. supply at low frequencies
is clearly fairly high, since it consists of the output impedance
of the smoothing equipment. This will vary greatly with
frequency, and from receiver to receiver, but cannot be small
unless the last condenser C be impracticably large.
Obviously, therefore, the A.C. output of the last valve will
set up a voltage across it which will be communicated directly
to the preceding stage and so cause regeneration or degeneration
according to the phase of the feedback. If the phase be correct
for regeneration, the bass response of the receiver will be
increased, and if the feed-back be sufficient, motor-boating will
occur. If the phase be such as to produce degeneration, there
47
WIRELESS SERVICING MANUAL
will be no motor-boating but the bass response of the apparatus
will be seriously reduced. The common expedient of curing
motor-boating by reversing the connections to the winding of
an A.F. transformer must be considered bad practice, therefore,
since by changing the phase relationships it merely converts
excessive regeneration into degeneration and does nothing to
prevent the feed-back.
The correct remedy for motor-boating is to introduce
adequate decoupling. Each A.F. stage except the last should
have its anode circuit decoupled by the usual resistance-
condenser combination. The addition of such decoupling to
the circuit of Fig. 28 is shown in Fig. 29, in which Ri is the
decoupling resistance and Ci the decoupling condenser.
It can be seen that the resistance and condenser really con-
stitute a simple filter circuit and they actually contribute
greatly to the smoothing of the H.T. supply and so to the
elimination of mains hum. The degree of decoupling obtained
increases with the values assigned to the components, and is
roughly proportional to the product of resistance and capacity.
The amount of decoupling required depends upon the degree
of amplification between the point which must be decoupled
and the anode circuit of the last valve, upon the frequency
Fif. 28 : A typical A.F. amplifier illustrating the way in which feed-back occurs
49
MOTOR-BOATING
response characteristic of the amplifier, and upon the impedance
of the H.T. supply. It is consequently impossible to give any
general ruling for circuit values.
Value of R C
Experience shows, however, that the RC product should not
be less than about 80,000 (R in ohms, C in microfarads) in an
arrangement such as that of Fig. 29 when the bass response is
good. If an additional A.F. stage were used, the RC product
for the first stage should not be less than 200,000, but that for
the second stage can still be 80,000. The value of the resistance
will naturally be as high as possible, since a high value resistance
costs no more than one of low value, whereas the price of con-
densers increases with capacity. The resistance value, however,
is limited by the permissible voltage drop across it, and this, of
course, depends upon the anode voltage required by the valve
and the H.T. supply available.
In a circuit such as that of Fig. 29, Ri is commonly about
20,000 ohms and Ci should then be some 4 f^F. In an earlier
stage, the valve does not usually need such a high anode voltage
and the decoupling resistance can then often be made as high
WIRELESS SERVICING MANUAL
as 50,000 ohms. The capacity of the condenser can then be
selected to provide the requisite decoupling, and it is seldom
that anything higher than 8 /xF is needed.
The use of a choke instead of a resistance for decoupling is
sometimes recommended and it has the advantage that the
voltage drop across it can be very small indeed. It is much
more expensive, however, and in the author’s experience not
very effective. To do any real good the choke should have an
inductance of several thousand henrys and this is obviously out
of the question. Moreover, there is a risk of resonance effects
occurring whereby the feed-back is increased by such a
“ decoupling ” circuit.
In the case of grid circuits, when cathode bias is used, feed-
back may occur from the anode to the grid circuit of the same
valve. Such feed-back is degenerative and does not cause
motor-boating, but reduces the stage gain. If the by-pass
condenser (C2, Fig. 29) is absent, there is little harm in the
feed-back beyond the loss of amplification, but if the condenser
is present but of too small capacity, there ^will be a reduction
in the bass response only.
To be effective the by-pass capacity must be very large,
and it is a safe rule to choose a condenser having a reactance
at the lowest important frequency of not more than one-tenth of
the value of the bias resistance. Thus if R2 is 1,000 ohms, C2
should have a reactance of about 100 ohms at the lowest fre-
quency considered important, usually 50 c/s. The nearest
standard value is 25 /^F, which has a reactance of 127*2 ohms
at this frequency.
Capacities of this
order are quite practi-
cable, for in most
cases the voltage drop
across R2 does not exceed
40 volts or so and may
be much less. Electro-
lytic condensers of 50 a^F
rated for working at 50
volts are quite reasonably
priced and entirely
suitable for use in such
circuits. For voltages
up to about 12 volts,
200 ^F condensers are
obtainable.
MOTOR-BOATING
fit. 31 : Grid circuit decoupling is ineffective v^ith resistance coupling
An alternative arrangement possible in some cases is to
adopt grid circuit decoupling as shown in Fig. 30. The bias
resistance is R2 and A.C. potentials developed across it are
prevented from reaching the grid by the decoupling system
Ri C. The values are in no way critical and it is nearly always
satisfactory to give Ri a value of 100,000 ohms and C a capacity
of I /iF. It is important to remember that C must not be an
electrolytic condenser. There is always some leakage in an
electrolytic condenser, and in this instance it would have the
effect of very appreciably reducing the grid bias actually applied
to the valve.
It should be remembered also that when cathode bias is
used, grid decoupling is only effective when the intervalve
coupling is by means of a transformer ; it is useless to employ
it with resistance or auto-transformer coupling. This is clearly
seen from Fig. 31 ; although potentials developed across R2
are prevented by the decoupling components Ri Ci, from
reaching the grid of the valve through the grid leak R3, they
can easily reach the valve through the alternative path provided
by the anode circuit decoupling condenser C3 of the previous
valve and the intervalve coupling condenser C2.
In such cases there is no alternative to the use of a large
51
WIRELESS SERVICING MANUAL
capacity bias resistance by-pass condenser as long as cathode
bias i^ retained, and for A.F. stages in general it is usually
the best system. When grid bias is obtained by means of a
resistance in the main H.T. negative lead which is common to
all valves, however, grid circuit decoupling can be employed
whatever the nature of the intervalve coupling ; in fact, it
really demands this form of circuit isolation. In general, how-
ever, cathode bias is better, for each valve is then independent
of the others for its grid bias.
When resistance coupling is used, the feed-back caused by
the common impedance of the H.T. supply is positive and
negative on alternate stages. In general, phase shifts in the
couplings prevent one from making use of this fact to reduce
the decoupling needed, except in the penultimate stage. Here
it is not uncommon to find decoupling omitted.
Its omission not only results in a saving of components but
enables a somewhat greater undistorted output to be obtained
from the valve, since the voltage drop across a decoupling
resistance is avoided. The disadvantages are the loss of the
smoothing effect of a decoupling circuit and some reduction of
bass response at very low frequencies. This last can be of
negligible importance, for it is not difficult to push below
audibility the frequency at which the response begins to drop.
With resistance coupling between the penultimate and output
stages, the inclusion of decoupling at this point does not help
to reduce motor-boating. If the penultimate valve is a pentode,
however, feed-back to its screen, but not its anode, is regen-
erative, so that screen decoupling is important. A pentode A.F.
stage usually has very high resistance values and quite small
capacities consequently give adequate decoupling. A screen-feed
resistance, for instance, is often about 0-3 megohm, and a
capacity of 0*2 /xF or so is then sufficient.
There is little which need be said about the methods of
tracing A.F. feed-back, for its appearance in a set is invariably
due to inadequate decoupling. If the set has previously
functioned satisfactorily the trouble will almost certainly be due
to an open-circuited condenser or short-circuited resistance.
The latter is easily found by checking the resistance values or
measuring the voltage drop across the resistance. The former
can, of course, be checked by measuring the capacity of the
condensers concerned, but it*is usually easier to keep a spare
4 /xF paper condenser, fitted with flex leads terminating in
crocodile clips, and connect it in parallel with each condenser
in turn. Do not forget to discharge the condenser after it has
52
MOTOR-BOATING
been used and never discharge it by short-circuiting it, for this
may damage the condenser. Always discharge a condenser
gradually through a resistance of at least 10,000 ohms.
R.F. Instability and Motor-Boating
Turning now to the second type of motor-boating, caused by
R.F. instability, this normally occurs only in sets fitted with
A.V.C. and the action is as follows : — Self-oscillation results in
a large detector input from the locally generated oscillations
and this means that the A.V.C. circuit biases the controlled
valves negatively by quite a large amount. The instability
consequently ceases, the detector input falls and the A.V.C.
bias drops to zero. Oscillation then recommences and the
whole cycle of events is repeated indefinitely.
The audible result is practically indistinguishable from true
motor-boating, the frequency being determined by the time
constant of the A.V.C. system. The remedy, of course, is
entirely different and for the procedure to be adopted reference
should be made to the chapter dealing with R.F. instability.
It is usually possible to distinguish this form of motor-boating
from that due to true A.F. feed-back because it will disappear
when the set is tuned to a strong signal, for A.V.C. will then
bias back the controlled valves and make the set stable.
The third type of motor-boating is usually apparent only
when a strong signal is tuned in. It is due to A.F. feed-back
to the anode or screen circuit of one of the R.F. or I.F. valves.
Continuous motor-boating cannot occur, for the couplings will
not pass low frequencies, but when a carrier is present the
low-frequency potentials fed back can modulate it, and they
then pass through the couplings as a modulation on the carrier.
Modulation demands non-linearity in one of the valves, so that
two remedies are available — to prevent the feed-back or to
remove the non-linearity. The latter may be impossible,
however, for the non-linearity may actually be caused by the
feed-back if this is great.
Modulation by A.F. Feedback
It is clear, therefore, that for the avoidance of this effect the
anode and screen circuits of R.F. and I.F. valves should be
thoroughly decoupled, and values chosen for the decoupling
components which are effective at low frequencies. This is not
always easy, for the voltage and current requirements of many
R.F. type valves prohibit the use of high value decoupling
53
WIRELESS SERVICING MANUAL
resistances. Resistances of only a few thousand ohms arc of
little use unless the decoupling condensers are of the order of
50 ftF, and these are rather expensive in high voltage ratings.
Fortunately, the trouble is one which occurs quite rarely in
practice. It is usually present only when an exceptionally
good A.F. amplifier is used having a very good bass response.
The best remedy, and probably the cheapest in the long run, is
to use separate H.T. equipment for the post- and pre-detector
stages, but with care it is possible to overcome it by modifying
the A.F. couplings so that there is a cut-off below about 25 c/s.
This affects quality to a negligible degree. The use of a well-
balanced push-pull output stage greatly reduces any tendency
to this form of motor-boating. When it occurs in a set fitted
with push-pull A.F. amplification, therefore, a very probable
cause is that the amplifier is imperfectly balanced at the lowest
frequencies.
Labora-
provides
voltage as
measurements
54
CHAPTER 7
INSTABILITY IN R.F. AND I.F. STAGES
O NE of the commonest troubles encountered in a receiver,
whether new or old, is instability in the R.F. or I.F.
amplifier. It is invariably due to the presence of un-
wanted coupling between circuits separated by one or more
valves and the search for a cure inevitably means tracing the
coupling. Unwanted couplings may be due to a common
impedance ; they may be electro-magnetic between coils and
wiring ; or they may be electric, between components and
wiring, and in the valves themselves.
It cannot be too strongly emphasized that when instability is
present there must be an amount of coupling between different
circuits which is excessive for the degree of amplification.
Since it is always impossible completely to eliminate stray
couplings, there is an upper limit to the amplification obtainable
with stability. This may seem a truism, but it is all too often
overlooked.
In the case of instability in a straight set or in the signal-
frequency amplifier of a superheterodyne, much useful informa-
tion can be gained from the manner in which the performance
varies over the waveband. It will usually be found that the
instability occurs chiefly at the high frequency end of the
WIRELESS SERVICING MANUAL
Fig. 33 ; If chc wiring be carried out so that any lead is common to two tuned
circuits, instability is probable
waveband for two reasons : — first, the amplification is usually
at its greatest at this end of the tuning range, and secondly, the
coupling provided by stray capacities increases with frequency.
Sometimes, however, the reverse effect is obtained, and
although a set may be quite stable at high frequencies, self-
oscillation sets in which the tuning control is rotated to the
low-frequency end of the waveband. The trouble is then
invariably due to common impedance coupling and is usually
quite easy to trace. A typical arrangement is shown in Fig. 32,
and the way in which the tuned circuits are completed forms
one of the greatest danger points.
Each tuned circuit as a whole is indicated by heavy lines and
in the receiver the wiring completing each circuit should be
quite distinct from that involved in any other. Now in a gang
condenser, the rotors are all in connection with the frame, and
it is a temptation to earth the frame at a single point only. If
this be done the circuit becomes effectively that of Fig. 33 in
which the lead marked AB represents the earthing lead of the
gang condenser. It is clear that the resistance and inductance
of this lead form an impedance which is common to both tuned
circuits and which consequently couples them together. It
might be thought that the impedance of such a lead, which
rarely exceeds six inches or so in length, would be negligibly
56
INSTABILITY IN R.F. AND LF. STAGES
small. This is not the case, however, and it is quite sufficient
to cause severe instability in many cases.
When instability of this particular type is found, therefore,
the first thing to do is to investigate the wiring of the tuned
circuits to make sure that there are no common leads. In
some receivers, the gang condenser is not earthed directly, but
its contact with the chassis is relied upon for the connections
to the frame. This is usually quite satisfactory provided that
sound contact between condenser and chassis is obtained at
several different points well spaced along the condenser.
The sudden appearance of instability in a receiver which has
been previously working well can often be traced to a poor
contact between condenser and chassis owing to one of the
fixing bolts having become slack. An ohmmeter test of circuit
resistance will not reveal a defect of this nature, for the D.C.
circuit resistance will be a minute fraction of an ohm. In a
similar manner, a common return lead to a decoupling condenser
may cause instability, and it is a wise plan to wire a receiver
in such a way that no lead is included in more than one circuit
except where experiment shows that the use of a common
connection causes no harm.
When the instability occurs at the high frequency end of the
tuning range, however, it is rare for common impedance
coupling to be responsible, and it is more usually caused by
electromagnetic or electric coupling or both. The former
is unlikely to appear in a receiver which has previously been
satisfactory unless a poor connection has developed between a
coil screen and the chassis. It is quite possible in a new set,
however, even with apparently well-screened coils.
It is not always realised that screening is never perfect, and
that the conventional coil screen merely reduces the external
field of the coil to something like one-tv\'entieth of what it
would be without the screen. Of course, when coils are
individually screened the
coupling between them is
reduced much more, the
field inside the second
screen due to the first coil
being about of what
it would be if neither coil
were screened. These
figures are very rough and
depend greatly on the
size of the screens and the
c
Fig. 34 : In order to reduce stray coup
connections in a tuned circuit should
close together
57
WIRELESS SERVICING MANUAL
material of which they are made as well as its thickness. They
do show, however, that even with screened coils appreciable
coupling may exist.
Stray electromagnetic coupling may also exist in the wiring
if this be carried out in such a way that large loops are formed.
This is particularly important in the case of tuned circuits.
It is clear from Fig. 34 (a) that if the two leads between coil
and condenser are widely spaced, there will be an appreciable
field which will link with other circuits, whereas if the leads
are run close together as in (b) no such field can exist, for the
field surrounding each lead is cancelled by the equal and opposite
field surrounding the other.
Capacity Couplings
In spite of the dangers of electromagnetic coupling, stray
capacitive coupling is probably responsible for 90 per cent, of
instability troubles. This can occur between any unscreened
components or wires, and it is naturally most important when
the difference of R.F. potential between the circuits is large.
It is necessary, therefore, to keep the grid and anode leads of
a valve well separated, and screening is usually needed. It is,
however, much more important to keep the anode lead of the
second stage away from the grid lead of the first, for as the
amplification between these two points is much greater, the
difference of R.F. potential is much larger and a smaller degree
of coupling will promote instability.
Keep Grid Leads Short
In order to reduce any tendency towards instability, one is
often advised to keep all grid leads as short as possible. The
idea behind this piece of advice is that, as the valve amplifies,
any feed-back to the grid circuit will be much more serious
than in the case of the anode circuit. It is, however, a scheme
which can be carried too far, for a little thought will show that
the anode lead of the first R.F. stage is just as much a danger
point as the grid lead of the second, their R.F. potentials being
in many cases identical.
An endeavour to obtain very short grid leads often means
abnormally long anode leads, with the result that one may
actually be worse off than if longer grid leads were employed.
In general, the most sensitive point is the grid lead of the first
valve, and then the aerial lead. Next in order, and of equal
importance, come the anode lead of the first valve and the grid
lead of the second stage and so on through the receiver.
58
INSTABILITY IN R.F. AND I.F. STAGES
With a new and unstable receiver, stabilisation resolves itself
into the removal of the stray couplings responsible for the
unwanted feed-back. This is not easy unless a systematic
method of progression be adopted, for degrees of instability are
difficult to determine and there may often be several different
kinds of feed-back present at the same time. Stability may be
achieved by the removal or reduction of them all, but not by
any one alone.
The best procedure is to modify the set in some way, so that
it is only just unstable, even if this entails a big drop in amplifica-
tion, for with the set in this condition the removal of any one
source of feed-back is likely to stabilise the set, and each source
of feed-back can be dealt with in turn. Thus, if a set be fitted
with a manual volume control of the type operating by changing
the grid bias of early stages, the set can readily be stabilised by
its adjustment. Modern receivers fitted with A.V.C. do not
normally include such a control, but it is usually a simple
matter temporarily to wire a variable resistance of some 5,000
ohms in the cathode lead of one of the valves. The resistance
should, of course, be shunted by a condenser of o*i /^F capacity.
The procedure is to set the control at such a value that the
set is only just oscillating, that is, so that the slightest increase
in the resistance value causes oscillation to cease. Changes in
the receiver can then be made and their effect upon stability
instantly determined. The wiring should first receive attention
and suspected leads can be moved with an insulated tool, such
Fig 35 : Instability may occur through coupling between such apparently dead
leads as AB and EF
59
WIRELESS SERVICING MANUAL
as a wooden pencil ; if the oscillation ceases on changing the
position of any wire, then it is obvious that the wire in question
is responsible for at least some of the feed-back. Its position
should be changed to one which is less dangerous or it should
be screened.
It is important to remember that so-called low-potential
leads can be nearly as dangerous as grid and anode wiring.
Referring to Fig. 35, it can be seen that for R.F. currents the
anode circuit of each valve comprises the lead from the valve
anode to the transformer primary, the transformer primary
itself, the lead from it to the decoupling condenser, and the
wiring between this component and the valve cathode. R.F.
currents flow through the whole of this path, and are no stronger
in any one part than in any other. Electromagnetic coupling
may thus occur between any part of the primary circuit of one
valve and any part of another. In particular, if the leads AB
or CD in Fig. 35 run close to and parallel with EF or GH,
instability is highly probable, although these are commonly
believed to be “ dead ** wires.
Sources of instability
When locating sources of instability it is necessary to try
each possible source in turn. The wiring is often responsible,
but inadequate decoupling of anode, grid, and screen-grid
circuits is commonly to blame. The anode circuits are usually
the most important and decoupling is generally necessary when
the receiver includes more than one R.F. or I.F. stage. The
grid circuits are next in importance and the screen circuits the
least likely to cause trouble. Decoupling of the screen circuits
is, in fact, usually unnecessary, and a simple by-pass condenser
often suffices. It is, however, always necessary to make sure
that the component used is in truth a condenser, for faulty
components are not unknown and an internal open-circuit can
be responsible for many obscure defects.
By-pass and decoupling condensers in R.F. or I.F. circuits
usually have a capacity of o*i /xF, except in cases where decoup-
ling must also be effective at low frequencies when the capacity
may be as high as 2-8 /xF. Decoupling resistances vary much
more ; where the aim is decoupling at R.F. only, values of
500-1,000 ohms are common, but where A.F. decoupling is
also considered the resistance may be 20,000 ohms in anode
circuits and 20,000 ohms to 2 megohms in grid circuits, the
high values being used in sets fitted with A.V.C. In many
cases, however, the resistance is chosen to give the correct
60
INSTABILITY IN R.F. AND LF. STAGES
anode voltage to the valve concerned, since its value is in no
way critical from the point of view of decoupling.
Feedback from the A.F. Circuits
Apart from defects in the purely R.F. circuits which can
lead to instability, the A.F. side of the receiver is often of
considerable importance. The detector should ideally be fed
with an R.F. or I.F. input and produce an A.F. output only.
In practice, it always gives an R.F. output as well as the wanted
A.F. voltages. Most receivers include a filter in the detector
output circuit for the purpose of confining R.F. currents to the
detector and of preventing them from being fed to the A.F.
amplifier. Now it should never be forgotten that no filter can
completely prevent the passage of the currents against which
it is discriminating. * However good the filter may be, there
is always some leakage ; in other words, the attenuation of
any practical filter is never infinite.
The filter in many receivers consists merely of an R.F. choke
and condenser as shown for a typical arrangement in Fig. 36.
When a choke is used, it is usually the self-capacity which is of
importance in the R.F. filtering and this may be as much as
3 /xftF — allowing for stray wiring capacities. For frequencies
higher than some 400 kc/s, the filter then reduces R.F. potentials
to about 3 per cent, of what they would be in its absence ;
this is for a capacity C having the usual value of o’oooi fiF,
In many cases, the
choke is replaced by
a resistance of 50,000
ohms or so, and the
condenser C usually
has a capacity of
0*0001 fiF although it
is sometimes as high
as 0*0005 fxF. At 465
kc/s, and using a resis-
tance of 50,000 ohms,
the reduction is to
about 6-7 per cent.
The resistance is
nearly as good as
the choke, and as it is
cheaper it is often
used.
Fig. 36 : A typical filter in the output of a
detector for preventing the leakage of R.F. or
I.F. currents into the A.F. circuits
61
WIRELESS SERVICING MANUAL
Cl CHOKE
Fig. 37 : The R.F. filtering with this detector circuit is not as good as that with
the arrangement of Fig. 38
It should be noted that the alternative arrangement of com-
ponents shown in Fig. 37 is not nearly as good, for the R.F.
input to the filter is very nearly the full voltage across the
tuned circuit, whereas with the arrangement of Fig. 36 only
the R.F. voltage set up across Ci is applied to the filter. As
regards the proportion of R.F. in the output, this circuit is
about ten to twenty times as good as that of Fig. 37 ; the latter,
however, can rarely be avoided in a straight set, for the tuning
arrangements demand one side of the tuned circuit being
earthed.
The method of Fig. 38 can, of course, be used in either
straight set or superheterodyne, and it has all the advantages
of the basic circuit of Fig. 36, being actually the same save for
62
INSTABILITY IN RJ. AND LF. STAGES
the point at which it is earthed and the point from which the
A.F. output is taken. It is often inconvenient to be unable to
earth the cathode of the valve, however, and it cannot be used
in a battery set unless the diode be replaced by its equivalent,
a metal rectifier, or one of the new indirectly heated duo-diodes
is used.
Now it can be seen that with the degree of filtering commonly
employed there is an appreciable R.F. or I.F. component in
the output of the filter. Any coupling to early stages may
consequently result in instability and this is particularly likely
to be the case when the R.F. or I.F. frequency is low and
resistance-coupled A.F. amplification is used. A good
resistance coupled amplifier will give appreciable gain at
no kc/s and it may pass to some extent frequencies as high as
465 kc/s. Unless the detector filtering be exceptionally good,
therefore, R.F. or I.F. currents may appear in the output
circuit of the A.F. amplifier in sufficient magnitude to cause
instability if fed back to the input.
It is usually important, therefore, to keep the loud-speaker
leads well away from the aerial and also from exposed parts of
the receiver, while the aerial should not be permitted to pass
close to the detector and A.F. circuits. It should not be
thought that the necessity for these precautions, which amply
demonstrates the presence of R.F. currents in the A.F. circuits,
is evidence of bad design however. The steps necessary for
almost perfect isolation
of the circuits are well
known but, unfortun-
ately, impracticable in
broadcast receivers. A
well-designed multi-
stage filter in the detector
output circuits and
separate complete screen-
ing of the R.F., detector,
and A.F. stages is really
necessary, but would
make the equipment ex-
pensive and cumber-
some.
Before concluding
these remarks on
instability, it may be as
well to mention a few
of the lesser known
Fig. 39 : In A.C. sett (o) condensers from
heater to chassis reduce couplings caused by
the heater wiring, and in battery sets (b), a con-
denser across the valve filament has the same
effect
63
WIRELESS SERVICING MANUAL
ways in which unwanted coupling may occur. It is well known
that, to be effective, screening must be properly earthed ; it is
not so generally known that if screening is not earthed it may in
certain circumstances not only be inefficient as a screen, but
actually introduce coupling. The heater and filament wiring
of a set may also be responsible for instability and where this
is suspected it is a good plan to connect by-pass condensers to
earth as shown in Fig. 39. The condensers should be mounted
as closely as possible to the valve concerned and have a value
of about 0*1 fJiF. In short-wave sets, however, o*oi /xF may
be more effective.
Even in A.C. sets it is now common to find one side of the
heater connected directly to the chassis as in a battery set,
instead of isolating both sides and earthing a centre-tap on the
heater supply. This rarely introduces hum, and has the great
advantage of making the heater wiring much more nearly dead
at radio-frequency. When it is adopted, condensers across the
heaters are rarely needed, for the live heater lead is fairly well
earthed to radio-frequency through the valve heaters them-
selves. They are sometimes necessary in short-wave equip-
ment, however.
In general, condensers are less likely to be needed across
valve heaters than across the filaments of battery valves. In
an indirectly heated valve, the cathode and heater are separate,
and the heater circuit only picks-up R.F. currents incidentally,
but in a directly heated valve the filament is the cathode and
necessarily carries R.F. current.
64
CHAPTER 8
FREQUENCY AND AMPLITUDE DISTORTION
T here are two main types of distortion which may be
encountered in receivers and amplifiers — frequency distor-
tion and amplitude distortion. Both are always present
to some degree, but in the highest grade apparatus they should
not be detectable even when a direct comparison with the
original can be made. Probably as much as 99 per cent, of the
apparatus in use to-day, however, distorts to an appreciable
degree.
The type of distortion most widespread is frequency
distortion and it is fortunate that the ear can tolerate, and
become accustomed to, quite a large amount, for interference-
free reception of anything but the local station would otherwise
be an impossibility. In the majority of receivers now sold
frequency distortion is deliberately introduced in order to
avoid interference, such as heterodyne whistles and sideband
splash, for there is no known method of eliminating this type
of interference without also removing the higher musical
frequencies from the wanted transmission.
Ideally, the whole range of frequencies from 30 c/s to
15,000 c/s should be evenly reproduced. In practice, fre-
quencies above 1 0,000 c/s are of little importance and it is
possible to restrict the response to about 8,000 c/s before any
really noticeable deterioration in quality takes place. If the
cut-off occurs at a lower frequency than this, however, the
change in quality becomes noticeable to all but the uncritical,
and as the cut-off frequency is lowered the reproduction at
first loses its brilliancy, then becomes lifeless, and finally
muffled, so that speech loses much of its intelligibility.
Few sets, of course, give a sharp cut-off at their nominal
upper limit of response and the falling-off at high frequencies
is usually gradual. The majority of small superheterodynes do
not give much output at frequencies higher than 4,000-5,000 c/s,
and when the tone control is in operation there is very little
output above 3,000 c/s, so that the reproduction can then be
only a travesty of the original.
Such sets, however, in spite of their defects from an idealistic
point of view, can give very acceptable results and most
listeners are satisfied with the performance. Many, indeed,
seem to prefer a severe cut in the top response, and it has
often been a matter of wonder at the number of people who
65
WIRELESS SERVICING MANUAL
operate a set with the tone control permanently in the top-cut
position when at its best the set is incapable of giving any real
high-frequency response ! The reason usually given for this
is that such people are incapable of appreciating musical quality.
The author does not agree with this, however, and in his view
there is often justification for the extreme top-cut so often used.
He believes that the tone control is “ turned down ” to reduce
the amplitude distortion which is so often prevalent in the less
expensive receivers. One result of amplitude distortion is to
introduce harmonics, and as they are of higher frequency than
the fundamental, they are reduced by cutting the treble response.
Three things combine to make the reproduction of many
small receivers unsatisfactory. The first, and least important,
is the sideband cutting in the I.F. amplifier, which restricts the
response to about 4,000-5,000 c/s. The second is a large peak
in the loud speaker response curve around 2,000-3,000 c/s.
The third is the large amount of amplitude distortion introduced
by the usual pentode output valve in conjunction with the
usual small output transformer, and also by some A.V.C.
systems.
A pentode is often operated to give 10 per cent. 3rd harmonic
distortion at full output. At low frequencies it may well give
more because the output transformer often has too small a
primary inductance to maintain the load on the valve. The
transformer itself can easily introduce 10 per cent, distortion,
and again 3rd harmonic, at low frequencies. A low primary
inductance can even result in a discontinuity in the valve
dynamic characteristic at low frequencies ; when this occurs
very high order harmonics are generated.
The peak in the speaker response — which sometimes reaches
20 db. — accentuates the percentage harmonic distortion in the
case of certain input frequencies. Thus, if the valve gives
2 per cent, distortion of a 1,000 c/s note, the 20 db. loud speaker
peak at 3,000 c/s will bring the audible distortion up to 20 per
cent. At 150 c/s too small a primary inductance in the trans-
former may result in a discontinuity in the characteristic.
There might then be, say, 0‘i per cent. 20th harmonic generated.
The speaker resonance would increase this to i per cent. —
quite an appreciable amount for such a high order harmonic.
The audible effect of the combination of amplitude distortion
and speaker resonance is to make the reproduction harsh, but
apparently brilliant. Reducing the high-frequency response by
the tone control partly offsets the effect of the speaker resonance,
but at the expense of nearly all frequencies higher than the
66
FREQUENCY AND AMPLITUDE DISTORTION
resonance, and by doing so it reduces the amplitude distortion
to the level of the valve and transformer alone.
The net effect is much smoother reproduction, and most
people prefer it. This does not mean, however, that they do
not appreciate a wide frequency response. It means merely
that they object to amplitude distortion much more than they
dislike the lack of the upper frequencies. The author has no
doubt that the full range of musical frequencies would be
appreciated if it could be obtained with a very low level of
amplitude distortion, with an absence of marked loud speaker
resonances, and without interference. Normally it is only in
local reception that it is possible to obtain the full frequency
response with freedom from interference, no matter how much
one is prepared to spend on apparatus.
In a set for general purpose use, the dictates of selectivity
limit the response to about 5,000 c/s. To make the best of this
limited range, it is necessary to have well-designed I.F. circuits
with a flat pass-band of some 10 kc/s and a very sharp cut-off.
The loud speaker must be free from any marked resonances,
and amplitude distortion must be kept to a very low level indeed.
In some of the older receivers, the reproduction can often be
greatly improved by fitting a more satisfactory loud speaker,
changing the A.V.C. system to a type which does not cause
amplitude distortion, and providing the output stage with
negative feed-back. This last sometimes entails the addition
of an extra A.F. stage.
These remarks must not be thought derogatory of the set
designers. Technique continually advances and results can
now be obtained which would have been impossible ten years
ago. Then the designer nearly always has a limit of cost within
which he must keep and do the best he can. The result is that
it is nearly always possible to improve a set if one is prepared
to spend money in doing so.
It is now necessary to consider those faults which may occur
in a set through a breakdown in some component and which
result in increased distortion. There are actually few which
can cause a loss of the higher musical frequencies, although a
short-circuit across the variable resistance of a tone control
circuit would certainly do so. Short-circuited turns in an A.F.
transformer may also affect the response at high frequencies,
but will usually do so more noticeably at low frequencies.
The response at low frequencies is much more likely to be
affected by defects which can develop in a receiver. An internal
open-circuit in the coupling condenser of a resistance-coupled
67
WIRELESS SERVICING MANUAL
Fig. 40 : A tone control can readily be fitted to any receiver by connecting
a variable resistance K and a condenser C across the coupling (a) or trans-
former primary (b)
Stage, for instance, results not only in a large drop in volume
but in a complete absence of the lower musical frequencies.
Open-circuits in anode circuit decoupling condensers also
affect the bass appreciably because feed-back is introduced.
The effect may be either an increase or a decrease in bass,
however, according to the phase of the feed-back.
An open-circuit in a bias resistance by-pass condenser does
not usually affect the frequency response, but only causes a
general drop in the amplification ; a reduction in the capacity
of the condenser, however, will certainly cause the low
frequencies to be attenuated.
As distinct from distortion which is a defect in a receiver, it
is often desired deliberately to introduce frequency distortion
of a particular kind in order to compensate for some defect or
to make the reproduction more suited to the acoustics of a
particular room. The ordinary tone control is one example of
this nature, and as usually fitted it enables the upper frequency
response to be reduced. If a receiver is not equipped with
such a control, it is an easy matter to fit one by connecting the
series combination of a variable resistance and a condenser
across an A.F. coupling as shown in Fig. 40 for resistance
coupling (a) and transformer coupling (6).
68
FREQUENCY AND AMPLITUDE DISTORTION
The values of the components depend not only upon the
range of control required but also upon the impedance of the
circuit across which they are connected. In general, a satis-
factory value for the resistance is 0*25 megohm, and the con-
denser should be found by trial for the range of control desired,
A common value for the condenser is o*oi fiF with transformer
coupling and for resistance coupling also when the coupling
resistance has a high value. When a low value of coupling
resistance is used, however, the capacity must be larger and
0*02 ftp may be needed. This value is also often suitable
when the tone-control is connected across the output trans-
former primary, but the resistance can then be reduced to
some 50,000 ohms.
An accentuation of the treble can be obtained if an inductance
is inserted in series with the condenser, and the resistance has
a fairly low value. The discrimination in favour of the higher
frequencies is only obtained at the expense of amplification,
however, so that varying the control resistance has a large
effect on the volume. In general, the inductance should be
about o* 1-0*2 H. and the resistance 1,000-5,000 ohms with
C = 4 /xF.
The tone control circuit of Fig. 40 may be used to accentuate
the bass response with no alteration except to the values of
components. The condenser C must then be large, o*i or
so, and the resistance low, 1,000-5,000 ohms. Here again, a
variation in the value of the resistance has a large effect on the
amplification.
It should be understood that both these accentuation circuits
must only be employed in the early stages of an amplifier
where the amplitudes are low, otherwise serious amplitude
distortion may occur. They both reduce the amplification by
at least the amount of compensation required, so that their
use will often necessitate the addition of an extra stage. Further
details of such circuits are given in Chapter 24.
Amplitude Distortion
The second type of distortion is also quite common and far
less tolerable. Amplitude distortion occurs through non-
linearity in some part of the receiver, usually a valve, but iron-
cored chokes and transformers can cause such distortion, and
it makes the reproduction of a harsh rasping character, par-
ticularly at large volume. When valves are correctly operated,
the degree of distortion which they introduce is quite small
provided that the input signal voltage does not exceed a certain
figure.
69
WIRELESS SERVICING MANUAL
The amount of distortion is measured by the amplitudes of
the harmonics relative to the fundamental frequency and is
expressed as a percentage. It is usual to take the overload
point of a triode as the one at which the second harmonic is
5 per cent, of the fundamental, and of a pentode when the
square root of the sum of the squares of the second and third
harmonics is lo per cent, of the fundamental.
This degree of distortion is usually considered tolerable, but
for the finest results the distortion in each stage should be kept
much lower, otherwise the total will be excessive. When com-
paring figures for the power output of valves, however, care
should always be taken to see that the figures are all for the
same degree of distortion, for the above definitions are not
always adhered to.
Check Voltages and Currents
Such points are of minor importance in service work, how-
ever, for the distortion which has to be remedied is usually a
defect which has occurred in previously good apparatus.
Where distortion is encountered, the first step should always be
to check over the voltages and currents of all valves, not for-
getting the filaments or heaters, for the commonest cause of
distortion is a defective valve, or a defect in the circuits supplying
a valve, which has resulted in incorrect voltages being applied
to it. Most distortion, therefore, will be automatically located
during the initial routine tests, and once the cause has been
found a remedy is usually easy.
It is useful to remember that with Class “A” amplifiers, dis-
tortion caused by a valve is nearly always accompanied by a
change in the anode current. Much information can be
gained, therefore, by connecting a milliammeter in the anode
circuit of the suspected valve and watching it when a signal is
applied. For complete freedom from distortion the meter
needle should remain perfectly steady, but in practice when
the volume is at a normal level the needle will kick slightly on
loud passages of music. Slight kicks are thus unimportant,
but even on loud passages, the change of anode current should
not be allowed to exceed about 5 per cent, of the standing
current.
With triode valves, when dealing with resistance-coupled
amplifiers or with transformer-coupled stages having grid-
circuit decoupling, it is often possible to determine the cause
of the distortion from the direction in which the meter needle
kicks. With a valve having a high resistance in its grid circuit,
70
FREQUENCY AND AMPLITUDE DISTORTION
an upwards kick of the meter needle indicates that the distortion
is occurring through anode bend rectification, that is, the valve
concerned has too low an anode voltage, too high a grid bias,
or too small an anode circuit load impedance.
On the other hand, if the meter needle kicks downwards, the
valve is passing grid current on the positive peaks of the signal,
and this is a sign that the grid bias is insufficient. When dealing
with pentodes, however, a decrease in the anode current does
not necessarily mean grid current, for it can also occur through
incorrect anode circuit conditions.
Resistance-coupled pentode intermediate stages are much
more likely to cause distortion than triode stages because their
operating conditions are more critical. The anode coupling
resistance, the screen-feed resistance and the cathode-bias
resistance are all closely inter-related, and it is usually very
important that the screen voltage should be lower than the
anode voltage. An anode coupling resistance of o*i--o*5
megohm is often adopted and the anode is only about 6o volts
positive. The screen must then be at about 20-30 volts only
and accordingly the screen-feed resistance is about 0*3-2
megohms. With such low voltages the currents are low, and
a high value cathode-bias resistance is needed to give the
necessary bias — it is often 2,000-5,000 ohms.
Any large increase or decrease of the resistance values,
especially the ones in the screen and cathode circuits, can result
in appalling distortion, for the working point is easily shifted
on to the bends in the characteristic. On account of the high
values of resistance voltage checks are not always of much
help, for the voltmeter itself draws too much current. The
best way is to check the resistance values with the ohmmeter,
and then to measure the currents with the milliammeter. The
actual voltages are then easily calculated.
Class AB ” and Class B ” Stages
As already stated most types of amplitude distortion occurring
in A.F. circuits are readily detected by the usual meter tests.
An exception occurs when dealing with Class “ AB ** and
Class “ B output stages, for with these it is normal for the
anode current to rise with a signal. The definitions of these
types of amplifier are given in Appendix I, and it will be seen
that each type can be sub-divided into two according to whether
or not it is driven into grid current. The subscript “ 2
indicates that grid current is permitted.
These systems are not often found in A.C. broadcast receivers.
71
W/RELESS SERVICING MANUAL
They occur chiefly in P.A. amplifiers, but they are likely to be
found in any equipment with an output exceeding about
1 5 watts. For broadcast use they are commonest in battery sets.
It should be understood that these systems are really some-
thing of a compromise between quality and economy of power
supply. They can only give perfect results when valves and
components are also perfect. This applies, of course, to any
arrangement, but the effects of small departures from perfection
are much more serious than is the case with Class A**
amplifiers.
The usual cause of distortion developing in battery-operated
Class “ AB ” and Class “ B amplifiers is nothing more
serious than an ageing H.T. battery. As a battery grows old,
not only does its voltage fall but its internal resistance rises,
and it is this increase of internal resistance which has such a
serious effect upon the quality of reproduction. With
Class “ A amplifiers it has little effect, for the average current
is constant, but with quiescent output stages resistance in the
H.T. supply causes the H.T. voltage to fluctuate in sympathy
with the anode current variations, whereas it should remain
constant. An H.T. battery which may still be good enough
for a satisfactory performance with a Class “A** amplifier
may thus have to be discarded when a quiescent type of output
stage is employed.
Parasitic Oscillation
Apart from the H.T. supply, the only defect which is likely
to develop in such amplifiers is greater deterioration in one
valve than the other or in one-half of the output valve than
the other where a double valve is used. When distortion is
very severe, one should always check the anode currents of the
two sides of the output stage. Class “ B *■ valves are often two
triodes, and Q.P.P. (or Class AB ”) valves two pentodes,
built into a single bulb, and the filaments of the two valves are
connected in parallel. It is, therefore, quite possible for one
filament to break without the other, and the receiver will
continue to function in spite of this, but there will be severe
distortion.
A check of the anode currents is the easiest method of
testing for this, and if it be found that one side passes no
current it is certain that one filament is broken, provided that
everything else is in order.
Excessive grid bias is another cause of distortion with such
valves, and in the interests of quality it is inadvisable to make
72
FREQUENCY AND AMPLITUDE DISTORTION
the initial anode current very small. It is often thought that
this quiescent current can be as low as i-z mA, but in general
a considerable improvement in quality is evident when the
grid bias is reduced slightly so that it is some 3-5 mA. The
improvement is particularly noticeable at low volume.
One of the greatest bugbears of quiescent output stages is
parasitic oscillation. It is difficult to trace because it may not
be present in the quiescent condition but only on the peaks of
the signal, and it is difficult to cure because many of the usual
remedies are inapplicable. Anode circuit stopping resistances
cannot be used because of the fluctuating anode current, but
with a Class “ ABi or Class “ Bi ’’ stage it is permissible to
employ stopping resistances in the grid circuit. It has been
found, however, that resistances in each grid lead are not always
as effective as a single resistance in the lead to the centre-tap
on the input transformer as shown at R in Fig. 41 .
The value of this resistance is not critical and satisfactory
results are usually secured when it is about 150,000 ohms ; it
should not, in general, be lower than this figure. The by-pass
condensers C in the anode circuit are also advisable in most
cases, partly to prevent the load on the valve from rising to an
excessive degree at high frequencies but chiefly because they
also tend to prevent parasitic oscillation. They are usually
73
WIRELESS SERVICING MANUAL
given a value of 0*005 each, and if these condensers be
employed in conjunction with the grid circuit resistance R
little trouble from parasitic oscillation will usually be
experienced.
The difficulties with a Class “ AB2 or Class “ B2 ” output
stage are considerably greater, however, for since grid current
flows it is not permissible to connect resistances in series with
the grid circuit. It is customary to connect condensers of
about 0*005 /xF across each half of the output transformer
primary for the same reasons as in the case of a Class “ ABi **
or Class “ Bi ” stage ; these are indicated as C2 in Fig. 42,
which shows the driver and output stages. The centre-tap of
Fig. 42 : The connection of resistances and condensers across the driver trans-
former of a Class ** AB2” or Class *’ B2 ” stage is sometimes necessary
the driver transformer secondary must be joined directly to the
bias battery, if the valve is of the type requiring a negative grid
bias, or to — L.T. if it is not.
The resistances Ri and condensers Ci are usually omitted,
but if parasitic oscillation be present, their addition may effect
a cure. The condensers should be tried first and the optimum
value is usually about o*oi /xF, but really depends on the input
impedance of the output valve. Larger capacities are per-
missible with valves operating with zero grid bias than with
types requiring a negative bias.
74
FREQUENCY AND AMPLITUDE DISTORTION
Damping the Driver Transformer
If the addition of the condensers proves ineffective, the
resistances Ri should be tried also. Here, again, the value
depends upon the input impedance of the valve and a lower
value can be used with the zero bias type of valve than with the
negative bias kind. The resistances, moreover, absorb power
which must be supplied by the driver so that when they are
used, and particularly when they are of low value, a larger
driver valve or an increase in its anode voltage, or both, may be
necessary to prevent overloading in the driver stage. In general,
the resistances Ri should be about 5,000-10,000 ohms.
One factor which greatly tends towards the production of
parasitic oscillation is leakage inductance in the driver and
output transformers. Some leakage is always present, but
care should always be taken to select good quality components
in which the leakage inductance has been reduced to a minimum
by the proper sectionalization of the windings.
The Detector
Distortion is by no means confined to the A.F. stages, how-
ever, and the detector is a prolific source. Grid detectors have
a bad name for quality, but this is only justified when the
operating conditions are incorrect. Properly used, the grid
detector is very nearly free from distortion ; it is certainly
75
WIRELESS SERVICING MANUAL
much better than the anode bend detector and introduces no
more distortion than a good diode detector. The grid detector,
in fact, is nothing more than a diode detector and a triode A.F.
amplifier in which a single electrode — the grid — serves as - a
diode anode and as the control electrode of the amplifier.
For correct operation, therefore, the grid circuit must be
designed for detection and the anode circuit for amplification.
In order to obtain distortionless reproduction the signal input
should be large, the grid condenser Ci (Fig. 43) should not
exceed o-oooi /aF and the grid leak Ri should not be greater
than 0*25 megohm. In a mains set, the grid leak must be
returned to cathode, but in a battery set it should normally be
taken to positive L.T. To obtain distortionless amplification
in the triode action of the valve, the signal input should be
small, the anode circuit load impedance high and the anode
voltage high.
It will be seen that one of the requirements for distortionless
detection in the grid circuit is just the opposite of one of those
for distortionless amplification, and this is the weak point of
the grid detector. If the input required for distortionless
detection is greater than the maximum for distortionless
amplification, distortion is inevitable, but if the converse is
true, then there need be very little, if any, distortion.
The minimum input for distortionless detection varies very
little from valve to valve and there is little which can be done
to alter it. The maximum input which the valve can accept
as an amplifier, however, is easily controlled and increases
with the anode voltage. A valve with a low amplification factor
is also advisable.
In general, practically perfect detection will be secured if
the anode voltage be adjusted so that, with no signal, the valve
passes an anode current of 6-10 mA, the higher the better,
and the signal input is of such a value that the current falls
by about 20 per cent, on the application of a signal. Both post-
and pre-detector volume controls are thus really necessary for
the highest quality together with a milliammeter permanently
wired in the detector anode circuit. The pre-detector control
should be used to adjust the detector input to the optimum
value for any station, as indicated by the meter, and the post-
detector control should be employed to adjust the volume to
the required level.
When used with a small signal input, the grid detector, in
common with both anode bend and diode detectors, will
introduce considerable distortion which may be as high as
76
FREQUENCY AND AMPLITUDE DISTORTION
25 per cent. A large input is always needed for distortionless
rectification, and the great advantage of the diode detector is
that there is no factor in the detector which limits the input
permissible.
Anode Bend Rectification
With anode bend rectification it is necessary to avoid any
flow of grid current if distortion is to be avoided, and the best
results are usually secured by employing a valve of moderate
internal resistance with a high anode voltage and a value of
grid bias sufficient to reduce the anode current to about o*i mA
with no signal. The anode current rises with a signal and,
just as in the case of the grid detector, there is an optimum
change of anode current for quality. It is hardly possible to
give even an indication of this optimum figure, however, for it
is dependent on so many factors which vary from set to set ;
it will, however, rarely exceed i mA.
The diode detector performs only one action , that of detecting,
and gives no amplification at all. In the ideal case it has no
overload point, but in practice an upper limit is sometimes set
to the input by the risk of a breakdown in the diode if this
voltage be exceeded. Most diodes, however, even the small
ones fitted to multiple valves, will withstand an input of 50 volts
or so safely, and some of the duo-diodes as much as 200 volts.
In the ordinary receiver, therefore, the diode detector input is
not critical, and the detector will not introduce appreciable
distortion as long as the input exceeds a certain minimum
figure. This lower limit varies with different diodes and their
operating conditions and
may be as small as i volt
or as high as 10 volts.
A typical diode de-
tector is shown in
Fig. 44. Although
practical circuits may
differ somewhat in de-
tail, particularly when
multiple-valves are used,
the connections are
always based on this
diagram. The require-
ments for correct opera-
tion are that the load
impedance to modula- ^ Th. diod. d««t6r i. universal
tion irequencies snail be in modem receivers
77
WIRELESS SERVICING MANUAL
high compared with the
diode impedance, and
roughly constant over
the whole range of
modulation frequencies,
while it shall not be
greatly less than the
D.C. resistance. The
load impedance at the
carrier frequency, how-
ever, should be small.
The load at very low
frequencies is provided
by the resistance Ri
which is also the D.C.
resistance of the load.
Maximum linearity is obtained when it has a high value, but at
high modulation frequencies it is shunted by Ci and Cz. The
reactance of the former represents the load at the carrier
frequency and must be as small as possible. The reactance of
Cz must also be kept small, otherwise the filter circuit is of
little benefit. There is consequently a maximum value for Ri
when the highest quality is required. Normal values for Ci
and Cz are o*oooi /xF with Ri about 0*25 megohm ; some-
times, however, the values are four or five times as large, but
there is then likely to be some loss of quality.
Even ignoring the shunting effect of the capacities,
however, the resistance Rz, which is normally the volume
control, can cause distortion, for it is effectively in
parallel with Ri as far as modulation frequencies are
concerned and the A.C. impedance of the load circuit is
less than its D.C. resistance. It can be shown that when
Rz is four times the value of Ri, distortionless rectification
can be secured for modulation depths up to about 80 per
cent, and this is quite good enough even for high quality
apparatus. Thus, when Ri is made 0*25 megohm, Rz
should not be less than i megohm. The coupling condenser
C3 is, of course, chosen in relation to Rz just as for a resistance-
coupled amplifier, and a normal value is o *01-0 *1 /xF.
Few defects are likely to arise in a circuit of this nature,
apart from open or short circuits. It should be remembered,
however, that if Rz changes its value so that it is no longer
large compared with Ri, distortion will be evident and will be
severe if Rz is much smaller than Ri . A short-circuit between
78
FREQUENCY AND AMPLITUDE DISTORTION
the slider and the upper end of R2 causes severe distortion at
low volume, but has no effect at full volume. One other fault
may be found and this lies in the diode itself. The normal
characteristic is of the form shown by the solid line of Fig. 45
but, should" the emission fail either through age or through an
overload, it assumes the form of the dotted curve and distortion
is likely to be severe with large detector inputs, while the
efficiency falls off.
Pre-Detector Stages
Turning now to the R.F. or I.F. side, serious distortion
may easily occur in these stages. This is not as well known
as it should be, for although the frequency distortion caused
by excessive selectivity is usually recognised, few realise that
amplitude distortion may be quite as great in these circuits as
in the A.F. stages.
Trouble rarely occurs when receiving weak signals, and it is
normally at its worst in local-station reception, and particularly
when the aerial system is good. If the set is not fitted with
A.V.C., the first valve is more likely to be responsible than the
last R.F. or I.F. amplifier, but it is not necessarily at fault,
since the trouble can arise in any stage. If the set has A.V.C.,
however, the last stage is usually responsible.
Variable-mu valves are, of course, practically essential when
volume is controlled, as it always is nowadays, by varying the
grid bias, and valves with long grid bases are often better than
those requiring only a small bias for anode current cut-off.
Whatever arrangement is used, however, it should be under-
stood that an overload point always exists ; there is a certain
signal strength which must be considered a maximum for that
set, and distortion, and often reduced volume, will be found if
a stronger station be tuned in.
Controlling the Input
With a good set containing no defect, such trouble is not
likely to be found at more than 10 miles or so from a local
station and then only when a particularly good aerial is used.
Two remedies of a simple nature are available. The first is to
fit a local-distance switch to reduce the input to the set for
local reception. There are many ways of doing this, but one
of the best is to fit a make-and-break switch to connect a
resistance between the aerial and earth terminals. The value
of the resistance should be found by trial, since the optimum
value depends on local conditions. It will, however, normally
be of the order of 100 ohms.
79
WIRELESS SERVICING MANUAL
The second remedy is better in that it involves no additional
control and it reduces any interference caused by the local
stations ; it is particularly valuable with superheterodynes, for
it greatly reduces any tendency to whistle production. It
consists of connecting a wavetrap in the aerial circuit and tuning
it precisely to the local station. If there are two local stations,
two wavetraps should be used.
There are, however, faults in the receiver which can cause
distortion in the reception of local stations. Defects of this
type which cannot be located by the usual voltage and current
tests are most likely to reside in the A.V.C. system.
Open-circuited resistances, short-circuited or leaky by-pass
condensers, or an open-circuited condenser feeding the A.V.C.
diode may be responsible. A defective A.V.C. diode itself,
however, may be the cause, and this is particularly likely to be
the case if the receiver functions normally on all but the local
stations.
It is easy to see that if the diode loses its emission, the set
will function normally provided that the signal voltages are not
so great as to cause the A.V.C. diode input to pass beyond the
straight part of the dotted curve of Fig. 45. On strong signals,
however, a limit is soon reached to the possible A.V.C. bias,
and overloading occurs in an early stage, whereas the bias
would continue to rise with a good diode having a characteristic
like that shown by the solid line.
Some A.V.C. systems inherently introduce amplitude distor-
tion. They are usually particular varieties of delayed diode
systems, and although they were considered satisfactory some
years ago they are not now to be recommended. The circuits
in question are dealt with in Chapter 16, and where one is found
to be causing distortion one of the more modern alternatives is
recommended.
90
CHAPTER 9
background noise and local interference
T he average listener finds noisy reception far more irritating
than many other faults, but there are so many possible
causes of noise, that the mere statement that a receiver is
noisy is of no help in diagnosis. It can be one of the easiest
of all defects to trace, but it can also be the most difficult ;
a cure, in fact, is not always possible.
The commonest cause of noise is a loose connection and it
is the easiest to trace, for in nearly every case it can be located
by tapping every part of the set in turn. The noise takes the
form of a characteristic crackle which is readily recognisable.
It should not be forgotten, however, that trouble of this nature
may not always lie in the receiver itself.
In a battery set, the H.T. and L.T. leads should be suspect,
and in a mains set the mains flex should not be overlooked.
The aerial and earth must always be remembered, particularly
when the noise is intermittent. The author has come across
many cases of noisy reception and intermittent changes of
volume being due to nothing more than a down-lead wire
rubbing against a gutter, or a bad connection at the lead-in tube.
Never forget to examine the earth lead. This is just as
important if the earth connection has been recently installed
as it is when it has been there for years. Gardeners seem to
consider an earth as a new variety of weed and treat it accord-
ingly ; if they do not uproot it entirely, they try the effect of
their pruning shears on the earth lead.
The difficult types of noise may be divided into two classes,
those which occur in the receiver and those which are external
in origin. The internal noise is usually in the form of a hiss
which appears whenever a signal is tuned in and may be of
great intensity. It is commonly known as superheterodyne
hiss, but it is by no means a monopoly of the superheterodyne,
and this name for it is really an undeserved libel on this excellent
receiver.
The limit is reached when the noise voltage on the grid of
the first valve, due to thermal agitation of the electrons in the
conductors forming its grid circuit, is of the same order as the
signal voltage. Actually, the signal must be rather stronger
than this, for the first valve also introduces some noise. At the
present time, the limit for a well-designed set is reached with a
sensitivity of about i ftV. That is to say, if a set has this
$1
WIRELESS SERVICING MANUAL
degree of sensitivity it will give standard output for a signal
producing i /xV between the aerial and earth terminals, but the
internal noise will, in practice, be inevitably of the same order.
It is, therefore, useless to attempt to receive weaker signals, for
they will be swamped by the set noise.
It should not be thought, however, that a set of high sen-
sitivity will give more noise on strong signals than one of low
sensitivity. It will not, for the two sets will have the same
sensitivity on that particular signal, since the amplification will
be reduced either by the A.V.C. system or by the manual
volume control. Actually, the highly sensitive receiver may be
slightly the quieter since it is not so likely to be operating near
a point of instability.
In any case, however, with a given set the background hiss
increases as the signal input to the set is reduced. It is, there-
fore, important to use as good an aerial as possible and to
couple it to the set efficiently.
These remarks apply to all sets, straight and superheterodyne,
and if both sets be properly designed there is nothing to choose
between them on the score of background hiss. In practice,
however, the superheterodyne is more liable to develop excessive
hiss than the straight set, and it is this excessive hiss which is
important, for it is as much a fault as a loose connection.
There is no doubt whatever that the commonest cause is a
defective valve, usually the frequency-changer, but it may be
caused in some degree by any defect in the pre-frequency-
changer circuits which has the effect of reducing the efficiency.
Next in order comes parasitic oscillation. This is usually un-
suspected and is unfortunately rather difficult to trace, but the
author believes it to be responsible for much background noise.
It can occur in R.F. or I.F. stages, but is more likely to be
found in the oscillator.
Parasitic R.F. Oscillation
When parasitic oscillation occurs in an R.F. or I.F. stage it
may usually be stopped by inserting suitable resistances in the
grid and anode circuits just as in the case of an output valve.
The resistances should be non-inductive and of low capacity —
the ordinary metallised or compound resistances are suitable —
and mounted as closely as possible to the grid and anode.
With care the leads to the resistances need not exceed a quarter
of an inch. The grid resistance can usually have a value of
100-500 ohms, and the anode resistance can be about 100-1,000
82
BACKGROUND NOISE AND LOCAL INTERFERENCE
ohms. If there are several stages of amplification it is some-
times helpful to use different values for the different stages.
Parasitic oscillation is much more likely in the oscillator, for
the operating conditions are already suitable for oscillation.
Parasitic oscillation really means oscillation at any undesired
frequency, but the term is usually employed to denote unwanted
oscillation at a frequency much higher than that of any circuit
apparently connected to the valve. E\ery wire in a set has
some inductance and has some capacity to other leads. All
sorts of unsuspected tuned circuits may be present, therefore,
and if they are suitably coupled oscillation may occur at very
high frequencies — 6o Mc/s and even higher. Such oscillation
is difficult to detect except by its adverse effect upon the per-
formance of the receiver.
The best method of detecting parasitic oscillation in the
oscillator of a superheterodyne is by checking the grid current.
Most modern frequency-changers use a grid leak type oscillator
with a grid leak of about 50,000 ohms, and the grid current is
about 0*1-0 *5 mA. If a meter having a maximum scale reading
of about I mA is available, it should be connected between the
grid leak and cathode, the positive terminal being joined to
cathode. The reading of the meter will depend upon the
amplitude of oscillation, the peak oscillator grid voltage being
nearly equal to i*2 times the product of the grid current (in
amperes) and the grid leak (in ohms).
As the tuning control
is rotated it will be found
that the current varies
somewhat, usually falling
at the lower frequencies,
for it is impracticable to
maintain the amplitude
of oscillation quite con-
stant over the waveband.
The change in current,
however, should be quite
smooth, and if it be
found at any point that
there is a sudden jump
either up or down in the
current, parasitic oscilla-
tion is almost a certainty.
If a suitable meter is
not available for this
Fig. 46 : The connection of a resistance R in
series with the reaction coil of an oscillator will
often cure parasitic oscillation
83
WIRELESS SERVICING MANUAL
test, it may be carried out with a milliammeter connected in
the oscillator anode circuit. With a grid leak oscillator the
anode current falls when the valve commences to oscillate, but
the magnitude of the change of current is not a reliable guide to
the amplitude of oscillation. A sudden kick of the meter
needle when the oscillator is tuned over its band, however,
can be taken as an indication of parasitic oscillation.
The remedies must usually be found by trial and error ; it is
often advantageous to rewind the reaction coil with resistance
wire, but as this may be inconvenient it is usually as satisfactory
to wire a resistance in series with it as shown in Fig. 46. Such
a resistance R should have a value of 100-5,000 ohms. It will,
of course, reduce the amplitude of oscillation at the correct
frequency, but this is not usually very important : certainly not
if it prevents parasitic oscillation. It may be remarked that
the choice of a suitable value will enable nearly constant ampli-
tude to be obtained throughout the waveband.
Local Interference
Perhaps the most commonly encountered noises, however,
are due to no defect in the receiver, but are of external origin
and are usually known as man-made static. It is usually easy
to distinguish the types, because set noises are normally in
the form of a hiss whereas external noise is generally less con-
tinuous and sharper in character, resembling a more or less
continuous barrage of atmospherics.
Such interference can be generated by any electric motor,
other than the synchronous and squirrel cage types, by ther-
mostats, generators, flashing signs, automatic telephones, and
internal combustion engines, to mention only the chief sources.
Whatever its source, the interference reaches the receiver in
one or more of three different ways — by direct radiation from
the source picked up by receiver or aerial, by conduction along
the supply mains, or by radiation from the supply mains.
The best method of avoiding such interference is nearly
always to suppress it at the source by connecting suitable
suppressors to the apparatus causing the interference. This is,
unfortunately, not always possible, for the sufferer is not always
the owner of the apparatus responsible for the trouble. If
tactfully approached, however, most owners will be willing to
discuss the matter and few will object to the necessary sup-
pressors being installed.
In cases where nothing can be done to eliminate the
interference at its source, however, it is necessary to consider
84
BACKGROUND NOISE AND LOCAL INTERFERENCE
alternative methods. When the interference is outside the
premises on which the receiver is operated the connection of a
filter to the supply mains at the point where they enter the
house is usually effective in reducing the interference, for it
greatly reduces both that portion which is conducted by the
house wiring and that portion which is radiated from it.
With D.C. mains, condensers connected to the mains as in
Fig. 47 (a) are usually satisfactory, but for A.C. supplies the
arrangement of (b) is to be advised. The condensers should
have capacities of about i mF and the fuses protecting them
should be rated at 2 amperes. The condensers themselves must
be of good quality and high voltage rating.
When fitting suppressors it is essential that the connecting
leads be very short, for even the small inductance of a few
inches of wire may seriously reduce the efficiency of the device.
In no case should the leads to the condensers exceed 6 or 8 in.
and the earth should be good.
Fig. 47 : The method of connecting filters to the mains is shown at (a) for D.C.
supplies and at (b) for A.C. mains
85
WIRELESS SERVICING MANUAL
In extremely bad cases of interference it may prove necessary
to supplement such a suppressor with choke coils. One coil
should be connected in series with each mains lead, usually
between the point at which the suppressor is joined and the
house wiring. Sometimes, however, the coils are better placed
between the suppressor and the main switch.
When the house wiring has thus been isolated from inter-
ference, any which remains is likely to be due to external
radiation. If the interference persists when the aerial is dis-
connected from the receiver, a simple filter should be tried
across the mains feed to the receiver itself. Such a filter can
consist of two o*oi fiF condensers, one being connected from
each side of the mains to earth. If the interference still persists,
the screening of the receiver is probably defective.
Belling-Lee
Interference
Suppressor
When silent operation has been secured with the aerial dis-
connected, any interference which occurs when it is in use is
obviously due to pick-up by the aerial and its lead-in them-
selves ; the ideal remedy is to employ an aerial which will not
pick up interference but which will still pick up signals. There
is, however, no such aerial, for the signal and interference are
both waves in space and the property of the aerial which enables
it to respond to the one must also let it respond to the other.
The only course, therefore, is to erect the aerial at a point
86
BACKGROUND NOISE AND LOCAL INTERFERENCE
where it is outside the field of the interference. Little or
nothing can be done with an indoor aerial, but an outdoor
aerial which is at a greater distance than lo ft. or so from any
part of the house is likely to pick up little interference except
from such serious sources as tramways or electric railways.
Although such an aerial may itself pick up little interference,
the lead-in, if of ordinary type, will certainly do so, for it will
pass through the field of interference. It is necessary, therefore,
to use a special form of lead-in to prevent such pick-up. The
simplest course is to use a screened lead-in with low-capacity
screened cable, the lead-in being screened the whole way from
the junction with the horizontal span of the aerial to the receiver
itself. This is not always convenient, however, for the overall
diameter of such cable is about J in. to i in., and in addition
to mechanical difficulties in its disposition, it may, in some cases,
be considered unsightly.
As an alternative, ordinary screened cable with an overall
diameter of less than J in. can be used with impedance matching
transformers at each end. Suitable transformers are obtainable
commercially, and one is mounted at the junction of the lead-
in with the aerial while the other is fitted at the receiver itself.
Some loss of efficiency is inevitable with all screened down-lead
systems, but this is not usually serious over the greater portion
of the waveband. In any case, it is a small price to pay for the
reduction of interference.
A screened lead-in introduces far too great a loss to be
satisfactory for short-wave reception unless proper R.F. cable
can be obtained. The alternative is to use a transposed
feeder as a lead-in. This may also be used on the medium and
long wavebands. As shown in Fig. 48, two separate lead-in
wires are used and are run about 2 in. apart, being transposed
every i to 2 ft. by using transposition blocks. Only one of the
wires is connected to the aerial, the other being left free at its
upper end.
With this system, the aerial winding on the first tuned circuit
in the receiver requires modification, for it must be earthed at
its centre-point and the two lead-in wires joined to the ends of
the winding. Ideally, the winding should be electrostatically
screened from the secondary, but this is not always necessary.
The operation of this system depends upon the fact that each
lead-in picks up the interference equally, but as they are
coupled to the tuned circuit equally and in opposite phase the
interference balances out, leaving only the signal picked up by
the aerial proper.
87
WIRELESS SERVICING MANUAL
Fig. 48 : An anti-interference system employing a transposed feeder for the lead-in
The use of a screened lead-in or transposed feeder is also the
only practical method of countering the interference caused by
tramways and railways, but it is not usually as effective as it is
in the elimination of other types of interference. It must be
remembered that these methods are only useful when it is
possible to erect the aerial itself at a point which is free from
interference, and this is not always possible when the trouble is
due to trams.
It is very often found that interference is caused by apparatus
on the same premises as the receiver, and although the screened
lead-in may be one remedy, it is usually better to silence the
gear responsible ; this is not only likely to be more effective,
but will also prevent interference in neighbouring sets. Inci-
dentally, the mains filter already described is of no use in this
case, and is unlikely to cause any reduction of interference in
the same house, although it may do so in neighbouring
premises.
The usual offenders are fans, hair dryers, vibrators, vacuum
cleaners, floor polishers, sewing machines, refrigerators, and
the like. Vacuum cleaners and floor polishers are not usually
BACKGROUND NOISE AND LOCAL INTERFERENCE
'
MOTOR MAINS C3:
: MOTOR
(a)
C4:
MOTOR -
FRAME (b;
MOTOR
FRAME
Fig. 49 : For suppressing the interference caused by small motors the arrangement of
(a) is often used with D.C. supplies, and that of (b) with A.C.
important, for they are generally used only for short periods
and in the morning. Fans and refrigerators are the most
important, for in hot weather they are likely to be in more or
less continuous operation. In general, the simple filters shown
in Fig. 49 will prove satisfactory ; the arrangement at (a) is for
D.C. mains and that at (b) for A.C. When the frame of the
motor is not earthed, Ci and Ca can be o*i /jlF, while for A.C.,
C3 should be o*i mF, with C4 only o-oi ^F. If C4 be made
larger, there is a risk of shock on touching the metal casing
of the motor.
Silencing Thermostats
Better suppression is likely to be obtained when the frame
is earthed, and this should always be done when possible even
if it means the use of 3 -pin plugs and a 3 -way cable in the case
of portable apparatus. When the frame is earthed, all the
condensers of Fig. 49 can have the same capacity of o*i /xF to
I mF.
In all cases, the suppressor unit should be mounted actually
on the apparatus itself so that the leads from the condensers
to the motor can be very short. In this connection, it may be
remarked that especially
convenient suppressor
units which are
designed for insertion
in the flexible leads are
commercially obtain-
able.
Such simple filters
often fail in the case
of refrigerators, for
THERMOSTAT CONTROL
X
Fig. 50 : A resistance-capacity circuit forms an
effective suppressor for a thermostat
89
WIRELESS SERVICING MANUAL
although they may silence the motor, they do not always reduce
the interference caused by the thermostat. The remedy is to
connect the series combination of a resistance and condenser
across the contacts of the thermostat as shown in Fig. 50. The
condenser should have a capacity of o*i /tF, while the resistance
should be 50-200 ohms ; it is often necessary to find the best
value of resistance experimentally, for it may be fairly critical.
In a book of this nature no attempt can be made to deal
in any detail with the many different cases which can arise in
practice. The commonest causes of interference and the
remedies have been indicated and to those wishing for fuller
information regarding the suppression of man-made static
three publications can confidently be recommended : “ British
Standards Specification for Components for Radio-Interference
Suppression Devices, Interference Suppression,”* and
‘‘Radio Interference Suppression’'* by Gordon W. Ingram,
B.Sc.
* The British Standards Institution, Publications Department, 28 Victoria Street,
London, S.W. i. Specification No. 613. 1940.
* Belling 8 c Lee, Ltd., Cambridf^e Arterial Road, Enfield, Middlesex.
* Electrical Review, Ltd., Dorset House, Stamford Street, London, S.E.i.
90
CHAPTER 10
THE ADJUSTMENT OF GANGING— STRAIGHT SETS
T he performance of a receiver depends very greatly upon
the accuracy of the ganging. Each tuned circuit must
be resonant to the incoming signal if the correct results
are to be secured, and if one or more is mistuned both sensitivity
and selectivity are bound to suffer. The design of circuits
which are capable of being accurately ganged offers many
problems which concern few but the set designer. The
adjustment of the ganging, however, is a relatively simple
matter and, if the receiver is in good condition and the proper
apparatus available, it need only occupy a few minutes.
The number .of tuned circuits commonly employed in a
straight set varies from three to six and, whatever signal fre-
quency the set may be tuned to, all circuits should be resonant
at the same frequency. In practice, this is attempted by using
tuning coils of nominally the same inductance and a ganged
condenser in which the sections are matched so that they are
all exactly alike. Trimmers are fitted primarily to permit the
equalisation of the stray circuit capacities.
Even when the ganged condenser is at minimum its capacity
is not zero, but some 15-30 /x/iF ; coils have a self-capacity of
7-12 m/^F, the valves themselves load the circuit with capacity
which may reach 15 and the wiring may easily add 5 m/^F.
The total effective capacity with the variable condenser at
minimum may thus total 50-70 m/zF, and may exceed this
unless care be taken in design.
These capacities restrict the tuning range of a receiver, for
with coils of a given inductance the highest frequency (lowest
wavelength) to which a receiver will tune is governed entirely
by the value of the stray capacities. The stray capacities also
govern the lowest frequency to which the set will tune, of
course, but they do not then have so great an effect for they
form a smaller proportion of the total capacity in circuit.
Now it is clear that the stray capacities are likely to differ
widely in different circuits, for in a given receiver one circuit
may have two valves connected to it while another has only
one ; one circuit may be connected to a valve of different type
from another, and so on. Trimming condensers are fitted to
the circuits, therefore, in order that the capacity of the low-
capacity circuits may be brought up to that of the highest so
that the capacity in all circuits will be the same.
91
WIRELESS SERVICING MANUAL
It is theoretically possible by correct design to secure perfect
ganging. In practice, it is not, for neither the coils nor the
ganged condenser are perfect. No attempt is usually made to
match coils more closely than it 0-25 per cent., and sometimes
the limit is as high as rb i per cent. The sections of the
ganged condenser are rarely matched more closely than ±0*5
per cent., and more often ± i per cent. Even in the best
conditions, therefore, the errors are likely to be as high as
it 0*75 to it 2 per cent., which represent degrees of mistuning
of some it 0*375 to zt i per cent. ; at 1,000 kc/s, as much as
3,750 to 10,000 c/s.
The best settings of the trimmers, therefore, are not neces-
sarily the ones which equalise the stray capacities but the ones
which lead to the best average results over the waveband.
The audible effect of the permissible ganging errors is by no
means as great as one might suppose, however, and the chief
thing to be guarded against is an excess of error. This occurs
chiefly in the ganged condenser.
Although its sections may be accurately matched when it is
new, they are unlikely to retain that matching after a long
period of use. It is not as fully realised as it should be that a
ganged condenser is quite a delicate piece of apparatus and
easily damaged. A jar which causes no visible defect may
appreciably affect the accuracy of the matching, and changes
may occur after a period owing to stresses in the metal of the
vanes. After adjusting the trimmers it is consequently neces-
sary to check the ganging to make sure that it holds reasonably
well throughout the waveband.
The ganging is also affected by regeneration, whether this
be due to deliberate reaction or to stray feed-back effects.
Regeneration is usually at its maximum when the set is working
at full sensitivity and, since in normal use the full selectivity is
then most needed, the ganging should be adjusted with the set
in this condition.
The Preliminary Steps
If the set has a manual pre-detector volume control, therefore,
it should be kept at maximum while ganging, and if it includes
A.V.C., only a weak signal should be used so that the detector
input is kept below the A.V.C. delay voltage and A.V.C. does
not function to reduce the sensitivity. This, of course, neces-
sitates the use of a proper output meter as a ganging indicator,
or else obliges reliance on the ear. If an output meter is not
available, however, it is usually better to use a larger input
92
THE ADJUSTMENT OF GANGING .
and a milliammeter or voltmeter as an indicator than to rely
upon the ear.
A test oscillator should be used and the low potential side of
its output connected to the earth terminal of the receiver, to
which the earth is, of course, left connected. The high potential
lead should really be screened, but this is not always very
important ; it must, however, be taken through either an
artificial aerial or a o*ooo2 mF condenser to a lead terminating
in a crocodile clip so that it may readily be connected to any
portion of the receiver without risk of short-circuit.
In order to gang a receiver on the medium waveband, set
the oscillator to 1,500 kc/s and clip its output lead to the grid
of the last R.F. valve. Set the tuning condenser in the receiver
at minimum and adjust the trimmer on the last gang condenser
section for maximum response, adjusting the output of the
oscillator to give a convenient indication on the output meter.
Then set the oscillator to 1,400 kc/s and tune the set to it
exactly by the main tuning control.
Leaving the tuning control at this setting, transfer the
oscillator clip to the grid of the preceding valve and adjust
the trimmer on the next section of the gang condenser for
maximum response. As the circuit comes into resonance the
output of the oscillator must be reduced, otherwise the output
will be too great to be read on the output meter. When one
circuit has been adjusted, transfer the clip lead to the preceding
stage and adjust the extra circuits which become operative.
In most cases, where only three or four tuned circuits are
used and particularly when the circuits are already roughly
tuned, it is unnecessary to go through the whole procedure.
After adjusting the last circuit in the manner described, the
output of the oscillator can be transferred directly to the aerial
terminal and, leaving the last trimmer alone, each of the others
can be adjusted in turn, working backwards towards the aerial,
for maximum response. When this has been done, it is usually
wise to check over the adjustment of all circuits, again starting
at the last circuit and working towards the aerial.
If the set has a calibrated tuning dial, the first step of adjusting
at 1,500 kc/s is unnecessary. Set the dial to read 1,400 kc/s
and the test oscillator to this same frequency. With its output
clipped to the grid of the last R.F. valve, adjust the last trimmer
for maximum output. Then transfer the clip to the grid of
the preceding stage and proceed as described above.
Ganging with Reaction
If the receiver is fitted with reaction, the trimming will be
93
WIRELESS SERVICING MANUAL
somewhat more difficult, for it is affected by the setting of the
reaction control. In general, reaction is needed only when
maximum sensitivity and selectivity are required, and it is
obvious that the ganging should be accurate under this condition.
After ganging in the manner described, therefore, the setting
of the reaction control should be increased while retrimming
the circuit to which reaction is applied, usually the last, until
the set is nearly but not quite oscillating.
It will be necessary, of course, to reduce the output of the
oscillator from time to time as the sensitivity increases to keep
the needle of the output meter on its scale. The trimming will
also become much more critical. When reaction has been set
fairly closely to the oscillation point, the early circuits should
be retrimmed and it will often be found that the optimum
settings are now more critical.
If too much reaction has been used, the set may break into
oscillation as the early trimmers approach their optimum
settings and if this occurs reaction must be reduced slightly
and the last circuit retrimmed.
This completes the ganging if the coils and condensers are
well matched, and there is nothing more to be done on this
waveband. The procedure on other bands is exactly the same
save for the frequency at which it is done. In the case of the
long waveband it is usually best to carry out the ganging with
the oscillator set to about 300 kc/s (1,000 metres).
Possible Defects
It may be remarked at this point that the purpose of the
initial adjustment of the last tuned circuit at 1,500 kc/s is
merely to ensure that the set is trimmed with the right capacity
in the trimmers to permit the set to be tuned to this frequency.
If this is not done, and all adjustments are made at 1,400 kc/s,
too much capacity may be used in the trimmers and the correct
waveband coverage will not then be secured. In some cases,
particularly when trying out a new set, it may be found that
no optimum setting for the last trimmer can be found even
with the ganged condenser at minimum. This is usually a
sign that the stray capacity on this circuit is excessive and an
endeavour should be made to reduce it.
If it occurs in a receiver which has previously been satis-
factory and which has a triode anode-bend or grid detector,
an open circuit in the anode-to-cathode by-pass condenser is
the most probable cause. The valve will then give appreciable
94
THE ADJUSTMENT OF GANGING
R.F. amplification and
have a very high input
capacity due to
anti - phase feed - back
through its grid-anode
capacity ; the input
capacity may be as high
as 0*0001 ftp, and it will
vary over the waveband
and so cause poor
ganging.
Even when full equip-
ment is not available,
ganging is in no way
difficult, but if a milliam-
meter is used instead of
a proper output meter
as an indicator of re-
sonance inasetequipped
with A.V.C., care must
be taken to connect it
correctly. It can be
connected in the anode
circuit of any valve con-
trolled from the A.V.C.
system, but it must be
connected at a point of
low R.F. potential. The
correct position for the
meter is shown at {a) in
Fig. 51 on the H.T.
side of the decoupling
resistance and the wrong
position at (6) where the
meter is between the
valve anode and the
intervalve coupling.
In this position the
meter will not only
increase the stray
circuit capacity and so
render correct ganging
impossible, but it
will probably cause
95
WIRELESS SERVICING MANUAL
instability. An alternative indicator which is often more con-
venient is a voltmeter connected across the bias resistance of one
of the controlled valves as shown at (r) in Fig. 51. The indi-
cation is not always as good as with a milliammeter, but it is
usually sufficient.
Ganging on a Signal
Where a test oscillator is not available ganging must of
necessity be carried out on signals and an output meter is then
of no value, for the modulation depth of the transmission is
constantly varying. A milliammeter or voltmeter as described
above can with advantage be used, however. The procedure
is practically the same as with an oscillator, but all circuits must
of necessity be operative the whole time. The trimmers
should first be set at what one would expect to be roughly the
correct capacities ; that is, the first and last trimmers for the
aerial and detector circuits will normally be nearly full un-
screwed while other trimmers will usually be set at about one-
half of their capacity. No difficulty should then be found in
tuning in some signal, and on this each trimmer can be roughly
adjusted.
Then tune in a station on as high a frequency (low wave-
length) as possible and adjust each trimmer in turn for maximum
response, starting with the one nearest the detector and working
backwards towards the aerial. Should it be found that for
any circuit there is no optimum, but that signal strength
increases as the trimmer is unscrewed, it is a sign that
ganging is being attempted with too little capacity in the
trimmers.
Each trimmer should be screwed up a little, therefore, the
station retuned at a lower dial setting, and ganging attempted
again. No more capacity than is necessary should be used in
the trimmers, of course, otherwise it may not be possible to
secure the correct waveband coverage. The treatment of any
reaction control is exactly the same as when an oscillator is
available.
Although various methods have at times been put forward
for checking the accuracy of ganging over the waveband, such
as metal spades which reduce the coil inductance when held
near them or ebonite plates which increase the condenser
capacity when inserted between the vanes, none are of much
use in the modern receiver, for coils and condensers are in-
variably screened. The only satisfactory check is the drastic
96
THE ADJUSTMENT OF GANGING
one of trying whether a better performance can be obtained at
a lower frequency by reganging at that frequency.
It is drastic, because it means that after the check the set
must be ganged again at a high frequency. The check, of
course, is not to see whether the ganging adjustments have
been properly performed, but to see whether the coils and
condenser sections are matched. If a check is made at a
number of frequencies and it is found that in one circuit the
trimmer capacity must be continually increased to maintain
resonance as the frequency is lowered, it is highly probable
that the inductance of the coil in that circuit is lower than the
inductance of the others.
Conversely, if the trimmer capacity needed continually falls
with a lowering of the frequency, the inductance is too high.
The same effects may be, but are not as likely to be, obtained
through mismatching of the ganged condenser sections. If,
however, the variations in trimmer capacity are somewhat
erratic, it is probably the condenser which is at fault.
If the trimmers be adjusted at 1,400 kc/s, for instance, it
may happen that one trimmer needs more capacity at 1,000
kc/s, but that this new value holds for still lower frequencies,
while at the lowest frequency possibly less capacity is needed.
Such an effect could not possibly be produced by mismatching
of the coils, and if the defects mentioned below are not present
one is consequently safe in concluding that the variable con-
denser is to blame.
In a case, such as a tuned-grid coupling, where an R.F. choke
is effectively in shunt with the tuned circuit, erratic tuning will
occur if the choke has a sub-resonance within the tuning range.
Such sub-resonances are liable to occur with sectionalised
chokes. Each section has a frequency of resonance with its
own self-capacity, and there are other such frequencies formed
by a combination of sections with the self-capacities. Air-core
chokes arc particularly prone to this effect because the coupling
between the sections is loose enough to permit individual
resonances. Iron-core chokes are usually free from it.
The aerial circuit, too, quite often tunes erratically because
of the effect of the aerial. This is especially the case on short
waves, for the aerial itself is resonant at multiples of its natural
frequency. The effect on the tuning of the first circuit, which
is coupled to it, is very similar to that of a multi-resonant choke.
There is no real remedy, and all one can do is to find the setting
for the trimmer which gives the best average results over the
band.
97
WIRELESS SERVICING MANUAL
Below about 1,500 kc/s, however, one can do something
because the usual aerial is then below its fundamental resonant
frequency. It is the method of coupling to the first circuit
which is then so important. Some of the most widely used
methods are fundamentally wrong.
Typical coupling arrangements are shown in Fig. 52. With
(a) the aerial coil is usually a few turns overwound on the earthy
end of the tuning coil — about 10 per cent, of the turns of the
tuned winding — while w^ith (6) the aerial tapping is at about
the same percentage of turns from the earthy end of the coil.
These tw^o circuits arc actually identical as far as the R.F.
performance is concerned, and they fail chiefly because the
fraction of the aerial capacity transferred to the secondary is
not a constant, but varies with frequency. It cannot, therefore,
be corrected at all frequencies by any adjustment of the trimmer.
The less widely used circuit of Fig. 52 (r) suffers from a
similar defect unless the coupling condenser Ci is so small
that the efficiency is very low^ A capacity of 25 /x/xF is often
used, but this is much too large for good ganging.
In addition to the ganging difficulties, these circuits also
tend tow’ards low efficiency at the low-frequency end of the
band and they reduce selectivity excessively at the high-
frequency end. They arc thus very much of a compromise
and consequently are now falling out of use.
There is, however, nothing wrong with the circuit of Fig. 52 (a).
If the aerial coil is made of much larger inductance than the
tuned coil, so that the aerial circuit by itself resonates well
below the tuning band, and it is coupled loosely to the tuned
winding, results are much better. Such an arrangement is
usually known as a “ large-primary ” tuning circuit.
The effect of the aerial is then to reduce the secondary
inductance by an amount which is nearly independent of
frequency and which can consequently be compensated by
increasing the inductance of the tuned winding. Usually a
small error of ganging occurs at the low-frequency end of the
band — below 600 kc/s on the medium waveband. In addition,
the efficiency is much more nearly constant over the band, and
the selectivity at the high-frequency end is improved.
Constructionally, this circuit is easily distinguished from
the “ small-primary ” type, because it is usually made with the
aerial coil wound as a narrow pile of many turns spaced one-half
inch or so from the tuned winding. Sometimes it is mounted
at the grid end of the tuning coil. There is then a small
capacity coupling between the aerial and the grid end of the
98
THE ADJUSTMENT OF GANGING
Fig. 52 : The way in which the aerial is coupled to the first tuned circuit
greatly affects the ganging of that circuit. The best results are usually obtained
with a “ large-primary ’* circuit. This often takes the form (a), but (c) shows
a convenient way of adding it to an existing set
coil. This makes the efficiency somewhat better at the high-
frequency end of the band, but the ganging is not quite so good.
This type of coupling can readily be applied to an existing
set by using a separate coupling transformer as shown in
Fig. 52 (d). The author has constructed transformers of this
type by winding 600 turns of No. 36 D.S.C. wire on a former
of 0-65 in. diameter — the coil being bunch wound to a length
of I in. A layer of paper over the outside provides insulation,
and the secondary Lz consists of 19 turns of No. 26 enamelled
wire as a single layer. An iron-dust core is used. The big
coil is used for Li of Fig. 52 (d) and the small one for L2 ; the
aerial should be connected to the inside of the big winding.
The total inductance of the tuned circuit is increased by
the connection of L2, but it is also reduced by the effect of the
aerial connected to the primary. The two effects are nearly
equal and as a result the transformer can be used with an
existing set without much inductance error.
This particular transformer was designed for a super-
heterodyne with an intermediate frequency of 450 kc/s, and
consequently has a bigger primary than would normally be
used with a straight set. The exact resonance frequency of
the primary circuit depends on the aerial capacity, but it is of
the order of 300 kc/s. For a straight set, one would normally
reduce the primary turns to about 400 and obtain slightly
higher efficiency.
99
WIRELESS SERVICING MANUAL
Mismatched Coils
When the coils are found to be mismatched and they are of
the iron-core type, the most probable cause is that one of the
cores has changed its position, whereas with an air-core coil
the only likely causes are loose turns or short-circuited turns.
This last will cause a very large change in inductance.
It is by no means easy to match coils accurately. The
measurement of inductance at high frequencies is a tedious
process and demands, at the least, a calibrated condenser, a
valve voltmeter, and an accurate heterodyne wavemeter and,
unless the greatest care be exercised in operating the gear, the
results cannot be relied upon to be better than ± i per cent,
or so.
The most suitable method of measuring inductance for
ordinary purposes, where absolute accuracy is unnecessary but
only relative accuracy from one coil to another, is undoubtedly
a low-frequency bridge. Such bridges are fairly expensive,
however.
Matching Ganged Condensers
If the ganged condenser sections are found to be mismatched,
and the condenser is of the type having split end vanes, it is
possible to rematch it while it is in the receiver by bending the
end vanes appropriately. The procedure is to set the con-
denser so that the first split section is about one-half enmeshed
as in Fig. 53 (a) and to gang the receiver accurately. Then set
the condenser so that the second split section is just about
fully enmeshed as in (b), adjust the oscillator to the
receiver and then tune each circuit exactly to the oscillator, not
by the trimmers, but by bending the vanes (2) on the con-
denser appropriately.
The operation is best
carried out with the aid
of an insulated rod, of
ebonite or wood, so that
hand-capacity effects do
not cause difficulty.
When the vanes have
been bent the right
amount the condenser
must be further rotated
so that t^he third
k F»f. 53 ; A fanf«d condenser can be rematched
‘ by bendinf the split end vanes appropriately
100
THE ADJUSTMENT OF GANGING
section is just en-
meshed, the oscillator
readjusted to the new
frequency and the pro-
cess repeated. This is
done for each seg-
ment, and then the
ganging readjusted in
the usual way by the
trimmers at a high
frequency.
The process is
tedious and not as easy
as it sounds, for the
vanes have so much
spring in them that it
is difficult to make
them stay in exactly
the desired positions.
Facility and speed
come only with
practice.
Although rare in this
country at the present
time, receivers are
sometimes found which
include inductance
trimmers as well as the
usual capacity trim-
mers, and it is naturally
possible to obtain more
accurate ganging with
such sets, for the coils
are automatically
matched exactly during
the process of ganging.
Two methods of in-
ductance trimming are
in use ; the first con-
sists of a copper disc
which can be moved
relative to the coil by a
screw adjustment, the
second is the provision
101
FIf. 54 ; For the maintenance of accurate ganging it it necessary that coupling and decoupling condensers be properly proportioned
WIRELESS SERVICING MANUAL
of some degree of movement to an iron-core, again usually by a
screw adjustment. Whichever be used, the method of ganging is
the same. On the medium waveband, ganging is first carried out
at 1 ,400 kc/s by the capacity trimmers in exactly the same way
as with an ordinary set. The oscillator is then set to about
600 kc/s, the set tuned to it, and the inductance trimmers
adjusted for maximum response, the actual process being the
same as if they were capacity trimmers. It is then necessary
to return to 1,400 kc/s and readjust the capacity trimmers.
For the most accurate ganging the process should be repeated
until no further adjustments at either frequency give greater
sensitivity. In practice, it will usually suffice if the trimming
be done tw^ice at 600 kc/s and three times at 1,400 kc/s. In
any case, the final adjustments should always be at the higher
frequency.
Other Sources of Inaccuracy
Although the chief causes of inaccurate ganging are mis-
matching of the coils or the ganged condenser, they are not the
only causes. An examination of a receiver for these other
causes should always be made, for while some sets are straight-
forward, others may contain several pitfalls for the unwary.
In mains receivers fitted with A.V.C. and in most battery sets,
the earthy ends of the tuning coils are returned to the earth
line through condensers in order that grid bias may be applied
to the valves — the general arrangement being as shown in
54 -
Consider the second circuit ; the condenser C2 is necessary
partly for decoupling, but chiefly to complete the tuned circuit,
which then consists of the coil tuned by the variable condenser
in series with C2. The presence of C2 thus reduces the total
capacity, for the capacity of two condensers in series is less
than that of either alone. It is
Fig. 55 : in a Alter coupled by mutual
inductance condensers must be inserted
if accurate ganging it to be secured
clear, therefore, that if ganging
is to be maintained every tuned
circuit in the receiver must
include such a condenser
whether it is necessary for other
purposes or not, and that the
capacities of all such condensers
must be the same.
Referring again to Fig. 54, it
will be seen that such a condenser,
C3, is inserted in the detector
102
THE ADJUSTMENT OF GANGING
tuned circuit in order to maintain accurate ganging and in this
circuit it performs no other function whatever.
If the aerial tuning system contained only a single circuit,
the condenser to be inserted in this would have the same value
as Cz or C3. Where a pair of coupled circuits is used, how-
ever, this does not always apply. For a filter coupled by
mutual inductance, the condensers of Fig. 55 should have the
same values as C2 and C3 of Fig. 54, but when a capacity
coupled filter is used as in this latter illustration the condenser
Ci must still have the same value as the others. The value of
Ci is decided by the filter coupling required, and for correct
ganging Cz and C3 must have the same capacity as Ci.
When testing a receiver, therefore, it is necessary to check
these condensers, particularly if the ganging does not hold.
An internal short-circuit in C3, for instance, would have no
other effect upon the performance of a receiver than to upset
the ganging and the symptoms would be almost indistinguish-
able from those associated with an excessive coil inductance in
this circuit.
Taylor Model 6SA Signal Generator
103
CHAPTER II
THE ADJUSTMENT OF GANGING—
SUPERHETERODYNE I.F. AMPLIFIERS
M ost receivers are now of the superheterodyne type
chiefly because of the higher degree of selectivity which
is obtainable with such sets. The adjustments are even
more important than in the case of straight sets, for incorrect
adjustments not only reduce the sensitivity and selectivity but
may cause whistles, which are much more noticeable under
average conditions. The trimming of a superheterodyne may
be divided mto two distinct categories — the ganging proper of
the signal-frequency and oscillator circuits, and the adjustments
in the I.F. amplifier. The latter must always be done before
the ganging and will consequently be treated first.
Basically, the trimming of the I.F. circuits is no different
from the adjustment of the ganging of a straight set, for a
number of circuits has to be tuned, each to the same frequency,
by means of trimmers ; it is, in fact, easier since the amplifier
functions at a fixed frequency and no considerations of main-
taining ganging over a waveband enter. The amplifier usually
consists of pairs of coupled coils separated by valves, each pair
of coils with its trimmers being termed a transformer. Most
amplifiers include two transformers, but some have three ;
in some cases the coupling between the coils is fixed, in others
it is adjustable by a sliding coil or pre-set coil, while in still
others it can be varied by means of a panel control for variable
selectivity.
The frequencies used for I.F. amplification vary greatly.
The most widely used are probably 465 kc/s, 456 kc/s, and
450 kc/s, but no kc/s, 125 kc/s, 473 kc/s, and 1,600 kc/s are
found in some of the older sets. At one time American sets
invariably had a frequency of 175 kc/s, but they now usually
have a frequency around 450-470 kc/s. It is most important
that the I.F. circuits should be adjusted to the correct frequency,
for the constants used in the signal-frequency and oscillator
circuits depend upon the intermediate frequency. Unless the
I.F. amplifier is correctly adjusted to the right frequency there-
fore, accurate ganging will not be secured.
It can be seen, therefore, that the use of a calibrated test
oscillator is much more important with a superheterodyne than
with a straight set. The adjustments can be made without it,
however, but they are much more difficult. When adjusting
KM
THE ADJUSTMENT OF GANGtNC
a superheterodyne the first step should be to find out the
correct intermediate frequency for that particular set ; it will
usually be given in the maker's service instructions or in some
of the literature describing the set.
The test oscillator should then be set to this frequency and,
if the amplifier is already approximately lined up, its output
should be connected between the grid of the frequency-changer
valve and the earth line, a fixed condenser or artificial aerial
being interposed in the high potential lead for insulation pur-
poses. The output meter should be connected up, and the
oscillator output adjusted so that a convenient deflection is
secured on the output meter. Each I.F. trimmer must then
be adjusted in turn for maximum response, starting with the
one on the circuit feeding the detector and working backwards
towards the frequency-changer.
If the circuits are widely out of adjustment to begin with,
the oscillator output lead should be connected first to the grid
of the last I.F. valve, and the trimmers in the last transformer
roughly adjusted. The oscillator lead can then be transferred
to the grid of the preceding valve and the trimmers on the next
transformer adjusted, and so on until the oscillator is connected
to the grid of the frequency-changer, when all circuits can be
accurately adjusted in the manner already described. As the
circuits come into resonance the output increases, and the
oscillator output will have to be reduced from time to time to
keep the needle of the output meter on its scale.
These adjustments are the basio ones to an amplifier in which
the coupling in the transformers is optimum or sub-optimum,
and when they have been done nothing further is necessary.
When a milliammeter or voltmeter is used as an indicator of
resonance instead of an output meter in a set fitted with A.V.C.,
however, the indications will vary according to the precise
arrangement of the A.V.C. system. When the A.V.C. system
is fed with the detector from the secondary of the last I.F.
transformer as in Fig. 56 («), the meter will indicate normally
during the trimming of both Ci and C2, just as if a true output
meter were used. If the A.V.C. system be fed from the primary
of the last I.F. transformer, however, as in (6), where C3 is
joined to the anode of the last I.F. valve, the indications are
different.
Trimming with a Meter
When a milliammeter or voltmeter is used as an indicator
its reading falls as the circuit being tuned comes into resonance,
m
WIRELESS SERVICING MANUAL
Fig. 56 : When the A.V.C. diode is fed from the last transformer secondary (o) a
miiiiammeter tuning indicator will indicate resonance for all circuits, but when
it Is fed from the primary (b) it will not indicate the trimming of C2 properly
and trimming is carried out for maximum change of reading
or minimum absolute reading. This holds for all circuits with
the arrangement of Fig. 56 (a), but only for the primary and
earlier circuits with (6).
Depending upon the degree of coupling used between the
106
THE ADJUSTMENT OF GANGING
coils in the transformer, trimming the secondary (C2) will have
no detectable effect upon the meter reading, or will cause it to
rise as the circuit comes into tune. This is brought about by
the fact that such a meter merely measures the effect of the
A.V.C. bias ; its reading depends upon the input to the A.V.C.
diode and not upon the detector input. With the circuit of
Fig. 56 (d) the secondary of the transformer is not really in the
chain leading to the A.V.C. system, for the tuned circuits
affecting this end at the primary.
If there were no interaction between coupled circuits, trim-
ming the secondary could thus have no effect upon the meter
reading. In practice, however, the secondary absorbs energy
from the primary and it absorbs most when it is tuned to
resonance. There is then a smaller voltage applied to the
A.V.C. system and the indicating meter rises. When trimming
a receiver of this type, therefore, it should be remembered that
in the case of the last circuit trimming must be carried out not
for minimum reading on the meter but for maximum. Where
no change in the meter reading is detectable, the ear must be
used as a judge of signal strength.
The use of a proper output meter is very desirable, for it
avoids difficulties such as the above. This circuit is by no
means an uncommon one ; on the contrary, it is probably more
widely used than any other since it tends to reduce sideband
screech.
In certain I.F. amplifiers, the coupling between the coils in
the I.F. transformers is fixed but it is in some cases greater
(a) - (b) (c)
Fig. 57 : The resonance curve of an I.F. amplifier with sub-optimum coupling
in the transformer takes the form shown at (a) whereas when tight coupling is
used it has a flat top (b). The resonance curve obtained when this type of
transformer is not properly trimmed is shown at (c)
107
WIRELESS SERVICING MANUAL
than optimum. This is done in order that a band-pass effect
may be obtained and the resonance curve shall approximate to
that of (b) in Fig. 57, instead of taking the form shown at {a ) . A
considerable improvement in the quality of reproduction
naturally results from the reduction in sideband cutting.
Unfortunately, it is extremely difficult to adjust such amplifiers
correctly without elaborate gear or temporarily modifying the
set, and the only result of the normal trimming procedure is to
produce a response curve of the type shown at(r). Such a
resonance curve is very unsatisfactory and causes both low
selectivity and distortion ; the selectivity, moreover, is greater
on one side of resonance than on the other.
The best course with an amplifier embodying transformers
of this type is to use a cathode-ray oscilloscope with a frequency-
modulated oscillator, for this apparatus permits a picture of the
resonance curve to be obtained. If this gear is not available
and the amplifier must be adjusted, the best course is to damp
the tuned circuits by connecting resistances across them while
making the adjustments. The coupling required to obtain a
double-humped curve depends on the resistance of the coils.
If this is increased sufficiently the resonance curve becomes
single-peaked and the circuits can be adjusted by trimming for
maximum output in the usual way. The subsequent removal
of the damping produces the desired double-peaked curve.
The requisite damping can be added either by increasing
the series R.F. resistances of the coils or by reducing the
dynamic resistances. The second course is usually much the
easier and is less likely to affect the reactances of the circuits.
It is done by connecting resistances across the coils while
trimming, and if the usual metallised or carbon-type resistances
are used, and connected carefully, they will affect the circuit
capacity by a negligible amount.
Two courses are possible. Each tuned circuit of a trans-
former can be damped simultaneously, trimmed in the usual
way, and have its resistances removed. Alternatively, one coil
only can be damped while the other is trimmed. This necessi-
tates moving the damping resistance from one circuit to the
other as the circuits are adjusted alternately. It is, therefore,
rather more troublesome, but has two great advantages ; the
circuit being adjusted is undamped and tunes with its normal
sharpness, and stray capacity to the damping resistance does
not affect its tuning.
It is, therefore^ the better scheme. When dealing with over-
coupled circuits connect a resistance across the transformer
108
THE ADJUSTMENT OF GANGING
primary circuit and adjust the secondary trimmer for maximum
output. Remove the resistance from the primary, and connect
it across the secondary ; then trim the primary for maximum
output. Repeat the process, trimming secondary and primary
with the resistance across the other circuit, until no improve-
ment can be obtained. Then remove the resistance entirely,
and by swinging, the test oscillator, check that a symmetrical
double-humped resonance curve has been obtained.
The value of the damping resistance required must be found
by experiment for it depends on the dynamic resistances of the
circuits and their coupling, both of which are usually unknown.
The value is not important, however, as long as it is low enough,
but if it is too low the sensitivity may be so reduced that an
excessive output from the test oscillator is needed. As a rough
guide, a resistance of the order of 10,000 ohms is usually
satisfactory.
Variable Coupling
Some receivers in which a double-humped type of resonance
curve is desired are fitted with transformers in which the
coupling between the primary and secondary coils can be
varied at will. The adjustment is then much easier, for the
transformers are presumably so designed that alterations to the
coupling do not affect the tuning. The first step is to loosen
the coupling so that by no chance can a double-peaked response
curve be obtained, and then to trim each circuit exactly for
maximum output in the manner already described. The
correct shape of resonance curve can then be obtained merely
by adjusting the couplings. In the absence of express instruc-
tions to the contrary, the best results will usually be obtained
when the coupling in the last transformer — the one coupling
the I.F. valve to the detector — is optimum. The coupling
adjustment should be moved, therefore, until the output meter
reads a maximum or until the loudest signal is obtained.
If the receiver includes only one other transformer, the
coupling in this one should be slightly tighter than the value
giving the greatest output. If there are two other transformers,
however, the first one, which couples the frequency-changer to
the first I.F. valve, should be set at just about optimum coupling,
while the second which couples the two I.F. valves is set at a
value appreciably greater than optimum.
These remarks regarding coupling apply whatever the inter-
mediate frequency as long as it is not higher than about 500
kc/s. With amplifiers tuned to 1,600 kc/s, the coupling should
109
+H.T, +H.T. +HT.
WIRELESS SERVICING MANUAL
never exceed opti-
mum, and in general,
should be appreciably
below it.
Some I.F. ampli-
fiers include variable
selectivity, the aim
being to permit the
resonance curve to be
varied within limits
by means of a panel
control so that the op-
timum compromise
between selectivity
and quality can be
secured under any
conditions. A typical
arrangement is shown
in Fig. 58, and it will
be seen that the
coupling between the
coils of two trans-
formers can be varied
w hile the coupling in
the third is fixed,
usually at a lower
value.
With loose coup-
ling, the resonance
curve is of the form
(a) of Fig. 59, repre-
senting high selec-
tivity. When the
coupling is increased
the response curve is
broadened (^) and the
selectivity is lower,
but there is less side-
band cutting. With
the full coupling the
resonance curve
should be essentially
flat-topped (c) and
the selectivity quite
THE ADJUSTMENT OF GANGING
low ; this condition is
usually only suitable for
high-quality local recep-
tion.
The adjustment of
such an amplifier hardly
differs from one with
fixed loosely - coupled
transformers. The first
step is to set the control
at maximum selectivity
and then to adjust the
trimmers for maximum
response in the manner
already described
in detail. It is more
Fig. 59 : The types of resonance curve obtainable
v^ith a variable-selectivity I.F. amplifier; (a) high,
(b) medium, (c) low selectivity
important than with a fixed-coupling amplifier, however,
to make sure that each circuit is tuned exactly to resonance.
for if any circuit is mistuned the correct fiat-topped resonance
curve will not be obtained at low selectivity.
When the trimming has been completed, therefore, set the
control to low selectivity and swing the oscillator over a range
of frequencies on either side of resonance. The output should
not vary by more than 2 to 3 db. over the range of frequencies
passed by the amplifier. If it should be found that the response
WIRELESS SERVICING MANUAL
of the amplifier is markedly asymmetrical, the most probable
causes are inaccurate trimming, stray capacity coupling between
the primary and secondary circuits, or excessive feed-back in
the amplifier, and the remedies are obvious.
Another type of variable-selectivity transformer is sometimes
used and the connections to one such transformer are shown
in Fig. 6o. The coupling between the coils remains fixed, and
the selectivity is varied by the variable resistance R, being
greatest when R is at maximum. To adjust a transformer of
this nature, set R at maximum and adjust Ci and C2 in the
normal way for maximum output. Then set R at minimum
and adjust C3 for minimum output.
The purpose of this third circuit is to absorb energy from
the main pair of coupled coils and the amount of absorption is
greatest when the resistance is lowest. The absorption is also
greatest at the frequency to which the circuit is tuned. When
R has a very high value, the circuit has little effect, but as R is
lowered the circuit absorbs more and more energy, and the
absorption increases more rapidly at resonance than at the
sideband frequencies, thus tending to give a flat-topped
Fig. 41 : An A.V.C. system which includes a circuit
tuned to the intermediate frequency
response curve.
Before concluding
the discussion of the
I.F. amplifier, men-
tion should be made
of a type of A.V.C.
system which may need
adjustment. The A.V.C.
diode is sometimes pro-
vided with a stage of
amplification of its own
which may be fed from
the grid of the last I.F.
valve, the anode of this
valve, or from the detec-
tor input. Typical con-
ncctionsfor this amplifier
are shown in Fig. 61 ;
sometimes a choke is
used instead of the
tuned circuit for the
coupling to the diode
and then no adjustment
is needed.
112
WIRELESS SERVICING MANUAL
Sometimes an ordinary I.F. transformer is used for the
coupling, and there are then two circuits to adjust. The
point of importance is that these circuits cannot be accurately
adjusted by means of an output meter, and it is necessary to
use a millimmeter or voltmeter connected to one of the valves
controlled by the A.V.C. system. The trimmer C can then be
accurately adjusted for minimum reading on this meter.
When no test oscillator is available, the adjustment of the
I.F. amplifier is more difficult, for the exact frequency is un-
known. The only course is by trial and error to get the various
circuits adjusted in such a way that some signal can be obtained.
The I.F. amplifier can then be adjusted in the manner already
described and the accuracy of adjustment will be no less than
if a test oscillator were used, save that the circuits may not be
tuned to the correct frequency. They will be in line with one
another, but unless the frequency is correct, the ganging will
not hold. It is, however, possible to determine the frequency
of an I.F. amplifier fairly accurately by methods which will be
dealt with later, and by a process of trial and error to arrive at
the correct frequency.
IN
CHAPTER 12
THE ADJUSTMENT OF GANGING—
SUPERHETERODYNE SIGNAL-FREQUENCY CIRCUITS
T he ganginpf of the signal-frequency circuits of a super-
heterodyne is basically the same as the ganging of a straight
set, but is complicated by the fact that there is an oscillator
circuit which must at all times be tuned, not to the signal-
frequency, but to a frequency different from that of the signal
by the intermediate frequency. In all modern superheterodynes
the oscillator frequency is higher than the signal frequency, so
that for the reception of a station on, say, i ,000 kc/s, the oscillator
is set at i,iio kc/s if the intermediate frequency is no kc/s,
and at 1,465 kc/s if the intermediate frequency is 465 kc/s.
Thus, while the signal-frequency circuits tune over the
range of 550-1,500 kc/s, the oscillator must tune over the
band of 660-1,610 kc/s for the low intermediate frequency
and over the band of 1,015-1,965 kc/s for the 465 kc/s inter-
mediate frequency. The constants of the oscillator tuning
circuit, therefore, are different from those of the signal-frequency
circuits, and their precise values depend upon the intermediate
frequency used.
There are two different methods of obtaining superheterodyne
ganging, but the one now almost universally employed is
known as the padding system. With this, the ganged condenser
is identical in every way with the type used in a straight set
and the difference in frequency between the signal and oscillator
circuits is obtained by padding the latter with a combination of
fixed condensers and by changing its coil inductance.
Just as in the case of
a straight set, the signal-
frequency circuits can be
represented by the com-
bination of Fig. 62 (a)
where L is the tuning
coil, C is the variable
condenser, and the
trimmer Ci represents
the total stray capacity.
The oscillator circuit is
shown at (b) and is similar
save for the introduction
of a padding condenser
Fig. 62 : The basic padding condenser circuit for
the oscillator is shown at (b) where C3 is the
padding or tracking condenser, and C2 is the
parallel trimmer. The signal-frequency circuit (a)
has only the trimmer Cl
115
WIRELESS SERVICING MANUAL
116
Fij. 63 : The signal-frequency and frequency-changer circuits of atypical superheterodyne showing the ganging arrangements
THE ADJUSTMENT OF GANGING
C3 in series with the variable condenser Co. (The
trimmer C2 is sometimes connected across Lo instead of
across Co.) It should be understood that at any dial setting
C and Co have the same capacity. Now at a high fre-
quency C3 has little effect upon the total cifcuit capacity, for it
is large compared with Co and C2 ; the correct oscillator
frequency is thus secured by the use of a lower value of induct-
ance for Lo than for L. At a low frequency, however, Co and
C2 together are too large to tune this circuit to the correct
frequency even with the reduced value of Lo, so that C3 is
included to reduce the total effective circuit capacity to the
correct value. This, in bare outline, is the principle of the
padding condenser method of ganging and those interested in
the precise mechanism or the calculation of the circuit constants
are referred to “ Ganging the Tuning Control of a Super-
heterodyne Receiver,” by A. L. M. Sowerby.*
It can be shown that in theory completely accurate ganging
is impossible with this system and that the best which can be
achieved is for it to be correct at three points in the waveband.
At all other points, some errors necessarily exist, the magnitude
of which depends upon the total width of the waveband and
the value of the intermediate frequency. In design it is the
general procedure to legislate for correct ganging at 1,400 kc/s
and 600 kc/s on the medium waveband and also at some arbitrary
frequency between these values which is usually 800-1,000 kc/s.
Now it is clear that in ganging a superheterodyne the aim is
to get all the signal-frequency circuits in resonance with one
another just as in a straight set, and by the adjustment of C2
and C3 to make the oscillator circuit tune always to the correct
higher frequency. The difficulties which arise in practice are
brought about by the fact that although it is the oscillator
circuit which must be padded to fit the signal-frequency circuits,
the tuning is governed entirely by the oscillator for it has the
whole selectivity of the I.F. amplifier behind it. Unless a
definite procedure be adopted, therefore, ganging may be quite
difficult.
There are several ways of tackling the problem and the one
adopted depends on circumstances. The one which the author
favours has been developed primarily for design purposes in
cases where for some reason circuit constants must be experi-
mentally determined, but it also forms the most satisfactory
method of adjustment when servicing a receiver. The great
advantage of the system is that it avoids any uncertainty ; each
• Wireless Engineer, February, 1932. S«e also Appendix I.
117
WIRELESS SERVICING MANUAL
adjustmeht can be made precisely and the simultaneous adjust-
ment of two circuits is avoided. In general, however, unless a
microammeter be available, the method is only conveniently
applicable to receivers which are fitted with A.V.C. and a test
oscillator capable of giving an output of up to about i volt is
also needed.
Consider the typical arrangement of Fig. 63. On the medium
waveband the switches Si, S2, S3, S4 and S5 are closed, but
S6 is open, whereas on the long waveband the first five switches
are open and S6 closed. The signal-frequency circuit ganging
arrangements are no different from those of a straight set and
consist merely of the trimmers across the gang condenser
sections which are adjusted on the medium waveband. In the
oscillator circuit, the trimming condenser C6 and the padding
condenser C8 have to be adjusted on the medium waveband
and the additional condensers C7 and C9 on the long waveband.
In some cases one of these last two condensers is omitted or is a
fixed condenser.
Now it is obvious that the best method of adjustment would
be to adjust the signal-frequency circuits on their own, and
then to set them at certain frequencies and bring the oscillator
into line by adjusting the appropriate trimming condensers.
This can only be done, however, if means can be devised where-
by the signal itself can be made to give an indication of its
presence without in any way necessitating the use of the
oscillator. It is possible to do this in a set fitted with A.V.C.
Suppose that the oscillator be prevented from functioning in
some way, such as by removing its anode voltage or by short-
circuiting its grid or anode coil with a short lead terminating in
crocodile clips. The application of a signal will then evoke no
response unless the signal has such a magnitude that it overloads
the R.F. or frequency-changer valves and they pass grid current
in consequence. In any set, the magnitude of the grid current
will depend upon the signal strength and it may, of course, be
read on a microammeter connected in the grid circuit.
A microammeter is not always available, however, and it is
an expensive and delicate instrument, so that it is fortunate
that in a set fitted with A.V.C. it is unnecessary. It is clear
from Fig. 63 that grid current in either of the two valves must
flow through R3 and R4. As a result of this current, a potential
is set up across these resistances of such polarity that the
A.V.C. line becomes negative with respect to the earth line
just as if the current through R4 were produced in the orthodox
fashion by the A.V.C. diode.
118
THE ADJUSTMENT OF GANGING
The potential on the A.V.C. line is naturally communicated
to the grids of all valves controlled from the A.V.C. system
with the result that their anode current falls. In the same
way that a milliammeter connected in the anode circuit, or a
voltmeter joined across the bias resistance, of one of the con-
trolled valves can be used as a tuning or ganging indicator, so
can such an instrument be employed in the ganging of the
signal-frequency circuits of a superheterodyne with the oscillator
inoperative.
The ganging of a superheterodyne is thus best carried out in
the following manner. Stop the oscillator from functioning by
short-circuiting one of its coils, connect a milliammeter or volt-
meter to one of the A.V.C. controlled valves in the usual way
to act as an indicator, set the variable condenser at minimum,
and connect the test oscillator to the aerial and earth terminals.
Set the oscillator to 1,500 kc/s and adjust it for full output ;
then tune to the signal thus applied by means of the signal-
frequency trimmers, C2, C4 in Fig. 63.
Even if the signal be modulated, no sound will be audible,
for the set oscillator is not functioning ; the tuning indicator
must be relied upon entirely to show resonance. When the
trimmers have been adjusted, set the oscillator to 1,400 kc/s,
tune the set to it by the main tuning control and readjust the
trimmers exactly. Should it be thought desirable, the accuracy
of the signal-frequency ganging can now be checked throughout
the waveband in the manner described for a straight set.
When the adjustment at 1,400 kc/s has been completed, it is
known that in the absence of any defect, the signal-frequency
circuits tune correctly and it only remains to line up the oscil-
lator circuit. Leaving the test oscillator at 1,400 kc/s and the
signal-frequency circuits tuned to it, reduce its output con-
siderably and make the set oscillator operative by removing the
short-circuit on its coils. Then adjust the parallel trimmer
(C6, Fig. 63) for maximum response, which may be read on an
output meter if the test oscillator be modulated, or on the
milliammeter or voltmeter previously used whether modulation
be employed or not.
The setting of this trimmer will normally be quite critical
and it may be found that there is more than one point at which
the signal occurs. This is particularly likely to be the case
with a low intermediate frequency, and especially if the trimmer
is of large capacity.
The reason is, of course, that there are two oscillator fre-
quencies which can convert any signal to the intermediate
119
WIRELESS SERVICING MANUAL
frequency, and if ganging is to be correct, it is essential to
choose the right one. In cases where two points of equal
response can be found, therefore, always choose the one which
requires the lower value of trimmer capacity.
It will sometimes happen that several response points are
found of different strengths. These occur through overloading
of one of the valves by the test signal and the output of the
oscillator should be reduced until only the correct response is
found. This correct point is always the strongest.
It is rare for any of these spurious trimming points to occur
on the medium waveband, but they are common on short-wave
bands and great care must always be exercised to select the
correct one.
Having adjusted the parallel trimmer, reapply the short-
circuit to the set oscillator, tune the test oscillator to 600 kc/s,
and adjust it for large output. Then tune the set to it by the
main tuning control, watching for the indication of resonance
on the milliammeter or voltmeter, reduce the oscillator output,
and remove the short-circuit from the set oscillator. The series
padding condenser (C8, Fig. 63) must now be adjusted for
maximum response in exactly the same way as the parallel
trimmer, choosing the lower capacity setting if two points of
response are found.
Since the adjustment of each trimmer in the oscillator circuit
affects the setting of the other to some degree, it is advisable to
readjust the parallel trimmer at 1,400 kc/s by repeating the
process already described for this frequency. It will rarely be
necessary to make any readjustment at 600 kc/s, however.
When this process has been carried out — and once the pro-
cedure has become familiar it can be done very rapidly — it will
be known that the ganging is as accurate as it can be made by
adjustments to the trimmers. There must of necessity be
discrepancies at some points in the tuning range, the magnitude
of which for the normal tuning range depends solely on the
intermediate frequency.
Nothing can be done to reduce these normal errors, but it
should not be forgotten that they will be greatly increased by
an incorrect ratio of signal to oscillator coil inductance or by
the use of an incorrect intermediate frequency.
On the long waveband, the adjustments are similar to those
on the medium waveband, save that there arc sometimes no
adjustments required by the signal-frequency circuits. When
the trimming condensers are as in Fig. 63, tune the set to the
test oscillator in the usual way with the set oscillator inoperative
120
THE ADJUSTMENT OF GANGING
and the test oscillator set to 300 kc/s ; then reduce the output,
remove the short-circuit from the set oscillator and adjust the
parallel trimmer (C7) for maximum response.
Then repeat the process at 160 kc/s, this time adjusting the
series padding condenser C9. If the receiver has no parallel
trimmer, only the series trimmer can be adjusted, and it is
usually best done at 160-200 kc/s. If the set has no adjustable
series padding condenser, but only a parallel trimmer, this is
best adjusted at about 250-300 kc/s.
The alternative and more usual ganging procedure does not
require that the set oscillator be periodically thrown out of
action, or that the set be fitted with A.V.C. It is, however,
somewhat more difficult to carry out and more likely to lead to
ambiguous results, although it is superficially easier. The pro-
cedure is to set the test oscillator to 1,400 kc/s with modulation
and connected to the aerial and earth terminals of the set.
An output meter is the best tuning indicator, but if the set
has A.V.C. , a milliammeter or voltmeter can be used. If the
receiver has a wavelength or frequency calibrated dial, set the
tuning control appropriately for 1,400 kc/s (= 214 metres) and
adjust the oscillator parallel trimmer (C6, Fig. 63) for maximum
response, and follow by adjusting the signal-frequency
trimmers.
If the receiver is not calibrated, it will be necessary to make a
guess at the appropriate trimmer settings, then to tune the set
to the signal by the main tuning control, and finally to adjust
the signal-frequency trimmers.
The test oscillator is then set at 600 kc/s and the set tuned
to it by the main tuning control. The series padding condenser
is then adjusted while rocking the gang condenser backwards
and forwards over a few degrees by the tuning control until the
optimum combination of settings is found. A return is then
made to i ,400 kc/s and the signal-frequency trimmers readjusted.
On the long waveband, the parallel oscillator trimmer should
be adjusted first at 300 kc/s while rocking the tuning control
until the best combination of settings is found. The series
padding condenser can then be adjusted in the same way but
at 160 kc/s.
Quite good results can be secured with this procedure,
particularly when it is merely a case of readjusting the ganging
of a set which has been in use for a time and which is already
presumably approximately ganged. In such cases, the adoption
of the more accurate procedure may be unnecessary, but with a
set which is widely out of adjustment it is to be recommended
K 121
WIRELESS SERVICING MANUAL
122
Fif. 6A : When a “ shaped-plate ” ganged condenser it used, the ganging arrangements are as shown here
THE ADJUSTMENT OF GANGING
as being more exact and probably more rapid because it avoids
any ambiguity. Great care should be taken to see that the
oscillator frequency is always higher than the signal-frequency.
Difficulty is unlikely in most cases where the series padding
condenser consists of a small capacity pre-set condenser in
parallel with a fixed condenser. Where a large capacity pre-set
condenser is used, however, and the intermediate frequency is
low, it is quite possible so to adjust the circuits that at 600 kc/s
the oscillator is higher than the signal-frequency, whereas at
1,400 kc/s it is lower.
The ganging cannot possibly hold over the waveband under
such circumstances and the performance of the set will be very
poor indeed in regard to sensitivity and its liability to second
channel interference, except at the two extremes of the wave-
band.
The Adjustments with a Superhet Condenser
I'he second system of ganging which is employed in some
of the older receivers necessitates no padding condenser for the
medium waveband. A special gang condenser is used, having
ordinary sections for the signal-frequency circuits but for the
oscillator a section in which the vanes are differently shaped so
that correct tracking is automatically secured.
The precise shape of the vanes depends upon the value of
the intermediate frequency, the stray circuit capacities, and the
relative value of the signal and oscillator circuit coil inductances,
as well as the tuning range. A condenser which has vanes cut
to suit an intermediate frequency of no kc/s, therefore, is un-
suitable for a frequency of 465 kc/s, and it would be unwise to
use it with a frequency of 120 kc/s, although the errors would
then naturally be smaller. On the long waveband the tracking
does not hold, and it is necessary to introduce a padding
condenser.
A typical circuit diagram is shown in Fig. 64 ; on the medium
waveband all switches are closed and ganging is best carried
out at 1 ,400 kc/s by adopting the procedure already described
for the adjustment of the padding system at this frequency. If
this procedure be adopted, which means setting the signal-
frequency circuits exactly to 1,400 kc/s with the set oscillator
inoperative and then trimming the oscillator condenser C6 ;
no other adjustments are needed on the medium waveband.
This method may not always be possible, however, and then
it is necessary to proceed rather differently. The oscillator
trimmer C6 should be set about half-way ; the test oscillator
123
WIREI£SS SERVICING MANUAL
should be at 1400 kc/s, and it should be tuned in by the main
tuning control, and the signal-frequency trimmers, C2, C4
adjusted for maximum response. The test oscillator should
then be set at 600 kc/s, and the set tuned to it. The oscillator
trimmer can then be adjusted while rocking the tuning control
backwards and forwards over a few degrees until the optimum
combination of settings is found.
A return must next be made to 1,400 kc/s and the signal-
frequency circuits readjusted as before. The oscillator trimmer
must then be readjusted at 600 kc/s and then the signal-frequency
circuits at 1,400 kc/s, and so on, until no further adjustment
at either frequency improves the performance.
On the long waveband, the ganging is no different from that
with the padding system, for it is actually the same method of
achieving tracking. It is not necessary, therefore, to treat this
case again.
The special gang condenser system of ganging is now rarely
used, for it is essentially a method for one waveband only,
although by the addition of padding it can be made to work on
two. Most modern sets have at least three wavebands and the
padding system is then much more suitable.
Although not commonly found, it may be as well to describe
the adjustments necessary in sets having inductance as well as
capacity trimmers. The large number of trimmers is apt to
be somewhat frightening at first, but if their adjustment is
tackled systematically, the presence of the inductance trimmers
will be found actually to help matters. This is because the
possibility of mismatched coils is removed ; it is always possible
to obtain correct ganging merely by adjusting the trimmers.
The procedure will be described in relation to the medium
waveband, but it is of course exactly the same on any other
waveband, the adjusting frequencies being proportionately
related to the limiting frequencies of the band.
A typical circuit is shown in Fig. 65 and it will be assumed
that it is necessary not only to gang the set but to adjust its
frequency coverage to a given band — say, 550-1,500 kc/s. It
is necessary to adopt the technique of adjusting the signal-
frequency circuits with the mixer driven into grid current, so
that a test oscillator with ample output is needed.
Short-circuit the set oscillator coil, set the tuning condenser
to minimum and the test oscillator to 1,500 kc/s. Using a large
output from the test oscillator and a meter to indicate resonance
in the manner previously described, adjust the signal-frequency
trimmers C2 and C4 for maximum response. Then set the
124
THE ADJUSTMENT OF GANGING
Fig. 65 : The most accurate ganging is obtained when the inductances are
adiustable as well as the circuit capacities. The detailed ganging drill for this
condition is given in the text
tuning condenser to maximum and the test oscillator to 550
kc/s, and adjust Li and L2 for maximum response.
Now go back to 1,500 kc/s and readjust C2 and C4. Repeat
the process, adjusting alternately at 550 kc/s and 1,500 kc/s
until no improvement can be secured. Adjust the capacity
trimmers only at 1,500 kc/s and the inductance trimmers only
at 550 kc/s, and finish by adjusting the capacity trimmers.
When this has been done the signal circuits are not only in
track with each other, but cover the required band of fre-
quencies. It is now necessary to gang the oscillator circuit to
the signal circuits, and one must remember that the parallel
trimmer C5 controls the tracking chiefly at the high-frequency
end of the band while the padder Cy and the inductance L3
control it at the low-frequency end and at the middle
respectively.
Set the test oscillator to 1,400 kc/s and tune the set to it by
the tuning control. Reduce the input to the set and remove the
short-circuit from the set oscillator. Adjust C5 for maximum
output. Replace the short-circuit on the set oscillator, increase
the test oscillator output and tune it to 970 kc/s. Tune the
signal circuits to it, reduce the test oscillator output, and remove
the short-circuit from the set oscillator. Adjust L3 for maxi-
mum output. Replace the short-circuit on the set oscillator,
increase the output of the test oscillator and set it to 600 kc/s.
125
WIRELESS SERVICING MANUAL
Tune the set to it, reduce its output and remove the short-
circuit from the set oscillator. Adjust Cy for maximum output.
The adjustments are now very roughly correct, but it is
necessary to repeat them several times to get them exactly
right for they are all interdependent. The best way now is to
leave L3 alone for the time being and to adjust C5 and Cy
alternately as previously described for a set with a fixed value
of inductance. Never forget that C5 must be adjusted only at
1,400 kc/s and C7 only at 600 kc/s, and always finish with C5.
When no further adjustments to these two improve matters the
tracking is correct at 1,400 kc/s and 600 kc/s. It is also correct
at one frequency between. What this frequency is depends on
the value of the inductance. There is an optimum value for
this frequency which makes the inevitable errors at other
frequencies a minimum. The next step, therefore, is to adjust
L3 correctly.
To do this, the test oscillator is set to 970 kc/s and the signal
circuits tuned to it in the usual way. The set oscillator is made
operative again, and L3 adjusted for maximum output. Then
leave L3 alone again, and readjust C5 and Cy at 1,400 kc/s and
600 kc/s alternately until they are again both correct at these
frequencies. Keep repeating the process of adjusting L3 at
970 kc/s, followed by C5 at 1,400 kc/s and Cy at 600 kc/s
alternately, and always ending with C5, until no further adjust-
ments improve matters.
The process is not nearly as difficult or lengthy as it sounds.
In general, for each adjustment of L3, it is not usually necessary
to adjust C5 more than three times and Cy more than twice.
It will also be unusual if L3 has to be adjusted more than three
times. If the set is not badly out of alignment to start with, of
course, many fewer adjustments will be needed.
Although not yet widely used in broadcast equipment,
inductance trimming will probably become much more common
in the future. Not only does it result in improved performance
because better ganging is secured, but it eliminates the necessity
for the accurate matching of coils in the factory. In these days
of iron-dust cores for coils, it is very easily arranged.
There is a further class of adjustments which, while hardly
ganging, are conveniently dealt with at this point. Some
receivers include a wavetrap in the aerial circuit tuned to the
intermediate frequency for the purpose of eliminating the
direct pick-up of signals of this frequency. A wavetrap of this
type is dignified by various names such as L.F. trap or I.F.
interference suppressor, but it invariably takes one of the two
126
THE ADJUSTMENT OF GANGING
Fig. 66 : Wavecraps in the aerial circuit are sometimes used to prevent the
break-through of signals on the intermediate frequency. A rejector is shov/n
at (a) and an acceptor at (b)
forms shown in Fig. 66, where the coil and condenser forming
the wavetrap are designated by L and C. Whichever circuit be
used, the adjustment is the same. In general, (a) is used with
“ small-primary ” aerial coils and (d) with “ large-primary
types.
The test oscillator should be connected to the aerial and
earth terminals and set to function at the intermediate fre-
quency, the output control being set so that some output from
the receiver is obtained. The trimming condenser C is then
adjusted for minimum output.
Some receivers, either as well or instead, include one or two
wavetraps of similar nature which are intended to be tuned to
the local station in order to prevent the formation of whistles
through the overloading of early valves by such strong signals.
These circuits can be adjusted by means of the test oscillator
in the same way as an I.F. trap, but using the appropriate
frequencies and tuning the set to the oscillator. In general,
however, it is better to perform this adjustment upon a signal,
the procedure being to tune the set to the local station and to
adjust the appropriate trap condenser for minimum response.
Before concluding, it may be as well to remark that the defects
which may occur in a superheterodyne and which prevent
127
W/RELESS SERVICING MANUAL
proper ganging are essentially the same as those which are
found in a straight set. Incorrect values of stray circuit capacity,
wrong values of decoupling condensers in A.V.C. circuits, bad
matching of the condenser sections, wrong coil inductance, all
apply with equal force. In addition, the use of the wrong value
of intermediate frequency will prevent proper ganging.
128
CHAPTER 13
TRACING THE CAUSE OF WHISTLES
A mong the various faults which may occur in a wireless
receiver, few are more troublesome to trace and cure
than those which give rise to whistles. The difficulty
in finding a cure which does not seriously affect the performance
in other ways is brought about by the fact that, while it is
generally easy enough to find out to which of the main types
any particular whistle belongs, it is by no means easy to de-
termine its precise cause.
As in every other trouble which may occur, diagnosis must
be carried out scientifically if it is to be accurate. The symptoms
will suggest a particular cause : the cause will suggest certain
tests which will confirm or disprove it ; the results of these
tests will suggest others, and so on.
The various whistles which may be met with are most
conveniently divided into categories according to the symptoms
which they produce. The three chief types are : —
(i) whistles which do not vary either in pitch or intensity
with the tuning ;
(2) whistles which vary only in intensity with the
tuning ;
(3) whistles which vary in both pitch and intensity
with the tuning.
In most cases, any given whistle can immediately be placed in
one of these categories, and the search for its cause corres-
pondingly limited.
Considering first the case of those whistles which are in-
dependent of tuning, they are almost invariably due to A.F.
regeneration . I f sufficient feed-back exists in the low-frequency
amplifier and it is in the right phase, self-oscillation will occur
at a frequency dependent upon the characteristics of the circuits.
When oscillation takes place at a very low frequency it is called
motor-boating, and this is treated in detail in another chapter.
At higher frequencies, however, the symptoms are in the
form of a continuous note which is usually of high enough
frequency to be called a whistle. The frequencies most usually
found lie between 1,000 c/s and 5,000 c/s, but in some cases
may be higher and even above audibility ; their presence may
not then be suspected.
With modern high-quality apparatus true whistles are usually
of fairly high frequency and due to capacity coupling between
129
WIRELESS SERVICING MANUAL
the input and output circuits. Consequently they rarely, if
ever, occur in apparatus in which the receiver and amplifier
are built on a single chassis, except when a pick-up or micro-
phone is being used, for the designer has taken precautions in
the construction against such feed-back. Where the amplifier
is a separate unit, however, feed-back is quite probable if the
various connecting leads are carelessly arranged.
The Loudspeaker Leads
The proximity of the loudspeaker leads to the input con-
nections, for instance, may easily cause instability, particularly
at very high frequencies and particularly if the amplifier has a
very good frequency response characteristic. Even if the
feed-back is insufficient to cause instability, it may affect the
characteristics of the amplifier.
Such feed-back can easily be cured by separating the leads
sufficiently, and where this is impracticable by screening them.
The screening should, of course, be earthed and when fitting
up an amplifier it is a wise plan to instal it at the start since it
also reduces the possibility of hum pick-up in the leads.
A whistle of this character is unlikely to develop with use
in a modern mains-operated receiver ; it is either present
initially or not at all. The case is different with a battery set,
however, particularly those of the older variety using A.F.
transformers of doubtful characteristics. Judged aurally, the
whistle is of similar character to that brought about by stray-
coupling between the input and output circuits, and may in
fact be actually due to this. More often, however, it is due to
nothing more than a partially run-down H.T. battery.
The internal resistance of an H.T. battery is quite low when
it is new, and little feed-back occurs between the different
stages even when decoupling circuits are omitted. When the
battery begins to age, but long before it is due for the scrap-
heap, its internal resistance rises and appreciable feed-back may
occur in the receiver. When actual instability results the
frequency of the whistle produced depends upon the character-
istics of the apparatus, and except when the frequency is low
the connection of a 2 /xF condenser across the H.T. battery
will often effect a cure.
Decoupling the Anode Circuits
The proper remedy, however, is to decouple the various
anode circuits. Feed-back of this nature is really the same as
that which produces motor-boating in a mains set, the different
130
TRACING THE CAUSE OF WHISTLES
symptoms being due mainly to the diflFerent nature of the
impedance of the H.T. supply. It is worthy of note that a
run-down grid bias battery may sometimes produce similar
effects, as also may a faulty battery cable or a dirty wander plug.
A whistle of comparatively low pitch, which is more
commonly called a howl, may occur through a microphonic
valve. The howl usually builds up slowly, and is often set up
by a loud passage of music. The offending valve is easily
located by tapping all the valves gently ; the noise resulting
from tapping the offender will leave one in no doubt whatever
that one has found the guilty specimen. If a non-microphonic
valve cannot be secured, the only remedy is to mount it in an
anti-microphonic valve holder or to mount the whole chassis
on sponge rubber.
A fault of similar character which occurs only on radio may
be due to a defective valve in the pre-detector circuits. In
general, however, the howl builds up very gradually and is of
quite low frequency ; moreover, it only occurs in selective
receivers and is more noticeable on the medium waveband
than on the long. It is then usually due to acoustic feed-back
from the loudspeaker to the vanes of the variable condenser
and it is a defect which is particularly likely to occur in short-
wave receivers.
The fault is one which is not always easy to cure when the
receiver and loudspeaker are in the same cabinet. The first
thing to do is to isolate the condenser from any vibration which
may be conveyed mechanically by the chassis and cabinet.
This may often be done by suspending the receiver chassis on
blocks of sponge rubber so that it can float freely.
Sometimes it is more convenient to mount the condenser on
the chassis in a similar manner, and sometimes both mountings
are necessary. The tuning dial must, of course, be arranged
to float with the condenser, otherwise the suspension is useless.
The effects of the direct radiation from the loudspeaker can
sometimes be nullified by enclosing the variable condenser in
a box stuffed with cotton wool, but this is often difficult to
arrange.
Vibrating Condenser Vanes
In extreme cases, it may be necessary to replace the variable
condenser by one having vanes less susceptible to vibration.
The materials used in the construction of the condenser have a
big effect upon its performance in this respect. It is often
stated that brass vanes are superior to aluminium from this point
131
WIRELESS SERVICING MANUAL
Fif. 67 : The introduction oft tuned whistle filter
ideally changes the response of the receiver from
the form of the solid line curve to that of the
dotted curve, but in practice the response is
more likely to take the form of the dash-line curve
I of view, but this is not
necessarily the case.
Actually, a condenser
v;^ -r — with vanes of soft
uj \ \ j / aluminium is likely to
z4 \ i j / be better than one in
2 \; :*/ which hard brass has
S \ 1 y been used, and the
'i-f author has obtained the
v; greatest freedom from
this form of feed-back
with condensers having
FREQUENCY zinc vanes. The re-
quirement is that the
Fif. 67 : The introduction oft tuned whistle filter vaneS be heavily damped
ideally changes the response of the receiver from
the form of the solid line curve to that of the SO tiiat tncy QO not
dotted curve, but in practice the response is vibrate easily, and this
more likely to take the form of the dash-line curve I’l 1 u
' IS most likely to be
found in condensers having vanes of soft metal.
The second class of whistles, those which vary in intensity
but not in pitch with the tuning, are rarely, if ever, due to the
receiver. These whistles are the beats formed between
adjacent stations and are inevitable if the receiver is capable of
reproducing the frequencies involved. Broadcasting stations
are usually spaced by 9 kc/s, so that if the receiver will reproduce
frequencies of this order, a 9,000 c/s whistle is inevitable.
High quality reproduction demands frequencies up to 10,000
c/s, so that trouble of this type is more likely to be met with in
modem high-fidelity apparatus than in older sets which do not
pretend to give a full high note response.
Whether or not such
a whistle is met with in r ■' ■
a high quality receiver | 1
depends entirely upon L I ...
the relative strengths of g ^ g
the stations involved. If oL C?S oL
both stations are of the o o
same strength the whistle |
will be prominent, xO I
whereas if the set is I I
tuned to the local station (a) (b)
and its neighbour is
quite weak, the whistle c- -ru # ci 1 «i •
^ ^ acceptor type of whittle filter it
may not be audible. shown at (0) and the rejector type at (b)
132
TRACING THE CAUSE OF WHISTLES
It is important to remember that it is impossible to prevent
such whistles from occurring without adversely affecting the
quality of reproduction. If a set does not reproduce them,
then it is certain that the receiver or loudspeaker is deficient in
the extreme treble. Conversely, if they are present, one has
the satisfaction of knowing that there is little wrong with the
high frequency response of the apparatus.
It is generally found, however, that the whistle is objected
to more than the slight deterioration ir quality which accom-
panies its removal. It is necessary to adopt the proper remedy,
however, if the effect on quality is not to be serious and most
emphatically the proper remedy is not to shunt intervalve
couplings with condensers nor to make use of the conventional
tone control. In order to remove a 9,000 kc/s whistle, an
attenuation at this frequency of at least 20-30 db. is needed,
and this cannot be introduced by such simple means without
attenuating seriously all frequencies down to about 3,000 c/s
and so making the reproduction intolerably muffled. The
proper remedy is to include either a correctly-designed low-pass
filter or, more simply, tuned rejection circuits.
The low-pass filter is not a general solution, for the filter
constants will vary somewhat with different receivers depending
upon the circuit impedances. Moreover, it is both bulky and
expensive. If it be used, it should have a cut-off frequency of
about 7,500 c/s and a frequency of infinite attenuation at
WIRELESS SERVICING MANUAL
9,000 c/3. For general
use, a tuned circuit is
likely to be of more
value ; this can be of
either the acceptor or
rejector type and is
really identical, save in
constants, with the
wavetrap commonly
used at signal frequency.
If the response curve
of the receiver is of the
form shown by the solid
curve of Fig. 67, then
the insertion of the trap
Fig. 70 : In an Intervalve circuit the trap should pimiif would ideallv
be connected in series v^ith the coupling condenser 7
change the response to
that shown by the dotted curve, introducing great attenuation
at the whistle frequency, but hardly affecting other frequencies.
In practice, the response obtained is more of the form of the
dash-line curve.
The two types of filter, acceptor and rejector, are shown in
Fig. 68 ; the acceptor circuit (a) is connected in parallel with an
intervalve coupling or output transformer, whereas the rejector
(b) is connected in series. The values assigned to the components
depend upon the frequency of the whistle, the amount of
attenuation required, the amount of attenuation permissible at
other frequencies, and the impedance of the circuits with which
the trap is used.
If each circuit is adjusted to give the same attenuation at
resonance and the coils are of the same efficiency, there is
nothing to choose between the two in the matter of performance.
For good attenuation a high Q (~ cuL/R) is needed, and this is
quite difficult to obtain, for the inductance required is so large
that small diameter wire must be used to keep the coil reasonably
compact. In order to secure adequate attenuation, therefore,
it is often necessary to permit more attenuation at other fre-
quencies than one would like.
In the case of the acceptor circuit, this means using a com-
paratively low value of inductance and large capacity, whereas
with the rejector circuit a high inductance and small capacity
is indicated. Now the capacity must be variable in order to
permit the circuit to be tuned exactly to the whistle and it is
easier to obtain a small variable condenser than a large one.
134
TRACING THE CAUSE OF WHISTLES
In general, therefore, the rejector circuit is the more useful
of the two.
The connections for such a rejector in the output circuit of a
typical diode detector are shown in Fig. 69, and in the case
of an intervalve circuit in Fig. 70. The value of the grid leak
resistance or volume control which follows the trap should be
about 0*25 megohm when the usual values for the trap constants
are employed. The filtering action is greatly increased if the
condenser Ci be included after the trap. The usual values
are L = o*8 H, C = 0-0005 /xF max., for a frequency of
9,000 c/s. For other frequencies both L and C should be
changed, and should be inversely proportional to frequency ;
thus, for 4,500 c/s, both inductance and capacity should be
doubled.
It may be remarked that the position of these components
in a set should be chosen with care, for as the coil contains
many turns it can easily pick up hum if it is too near a mains
transformer. It should, therefore, be kept well away from
any components which are likely to produce a stray magnetic
field.
i
Taylor Model 90A Multi-
Range Te»t Meter
135
CHAPTER 14
SUPERHETERODYNE WHISTLES
T he last and most important class of whistles contains all
those which vary in both pitch and intensity when the
setting of the tuning control is altered. The whistles
caused by instability actually come into this category, but this
trouble is dealt with in another chapter. There is rarely any
difficulty in deciding whether a receiver is unstable or not,
for instability gives other symptoms than whistles. The
whistles discussed in this chapter can occur only in super-
heterodynes and are commonly called second channel or image
interference, although this is actually only one out of many
possible causes.
The remedies for the different types of whistle which may
occur naturally vary with their cause, and it is necessary to be
able to distinguish them in order to apply the appropriate
remedy. This is not always easy, for the only audible difference
between some of the whistles is that the rate of change of pitch
for a given movement of the tuning control is different for some
whistles than it is for others, and in some cases there is no
difference at all. It is essential, therefore, to understand how
the whistles arise in order to apply the appropriate tests in-
telligently.
Second channel or image interference is the best known, but
not always the most important, source of whistles. It arises
in this manner. In order to receive a given station, the oscillator
is set to work at a frequency higher than that of the station by
the intermediate frequency. The intermediate frequency is,
in fact, the difference between the signal and oscillator fre-
quencies.
Normally, operation is carried on with the oscillator at a
higher frequency than the signal, for this leads to convenient
ganging, but it is not essential to the process of frequency-
changing, and the oscillator might equally well be lower than
the signal frequency. In actual fact, with the present crowded
condition of the ether, although the oscillator frequency is
higher than that of the wanted signal, there is always some
other signal at a frequency higher than that of the oscillator by
an amount roughly equal to the intermediate frequency.
Thus, suppose the intermediate frequency is no kc/s ; for
the reception of a station A on i,ooo kc/s, the oscillator will be
functioning at i,iio kc/s. Now if there is a station B on
136
SUPERHETERODYNE WHISTLES
1,220 kc/s, there may be interference. If both signals A and B
are handed on to the frequency-changer there certainly will be
interference, for the frequency-changer cannot distinguish
between them — both are separated from the oscillator by
no kc/s and both will be changed to this frequency and pass
through the I.F. amplifier. There will be no whistle, however,
for both stations produce a frequency of no kc/s exactly.
How a Whistle is Produced
Suppose the tuning to be changed slightly, however, so that
the oscillator is at i,ni kc/s. Station A will now produce an
intermediate frequency of in kc/s, and station B one of 109
kc/s. Both will still pass through the I.F. amplifier, but there
will now be a whistle of 2 kc/s. Thus mistuning the set by
I kc/s has produced a whistle of 2 kc/s, and it may be said that
the pitch of the whistle produced by second channel interference
changes its pitch for a given movement of the tuning control
twice as rapidly as does the whistle produced by an oscillating
detector set.
A whistle is produced if station B is not separated from
station A by exactly this amount. Suppose the oscillator is set
at 1,110 kc/s for the correct reception of A, and it produces
an intermediate frequency of 1 10 kc/s from this station. Then
if station B has a frequency of 1,218 kc/s, it will produce in
thefrequency-changerafrequency of 1,218 — i,iio= 108 kc/s.
This will pass through the I.F. amplifier along with the correct
frequency of no kc/s produced by A and in the detector a
beat of no — 108 = 2 kc/s will be produced.
There are many other possible causes of whistles, among
which the following are the most important : —
I.F. Harmonic Interference : This is similar to second
channel interference, but the interfering station is on a
frequency either higher or lower than that of the oscillator
by one-half or one-third of the intermediate frequency for
the second and third harmonics respectively. Higher
harmonics are not usually important. The harmonics
are usually generated in the frequency-changer (fx =
is -f fi ± ft In).
Beat Interference^ Fundamental : This occurs from a
station spaced from the wanted signal by an amount equal
to the intermediate frequency {ix *= f^ ± ft)-
Beat Interference, Harmonic : This is similar to the
foregoing but non-linearity in the frequency-changer is
137
WIRELESS SERVICING MANUAL
necessary for it to appear. It occurs from signals spaced
by one-half or one-third of the intermediate frequency
from the wanted station for the second and third harmonics
respectively (f^ = fs ± fi/n).
Oscillator Harmonics : This form of interference is
caused by harmonics of the receiver oscillator, usually the
second or third, beating with signals to produce the
intermediate frequency. This type of interference is
usually confined to the long waveband (fv = i f^).
In the foregoing, fx = frequency of station causing the
interference ; is = frequency of station upon which the
interference is found ; ii = intermediate frequency ; and
n = order of harmonic involved.
I.F. Harmonic Feed-back : This type of interference
occurs when harmonics of the intermediate frequency
generated in the I.F. amplifier or detector are fed back to
the pre-frequency-changer circuits.
With the degree of pre-selection included in most sets, all
forms of interference, other than I.F. Harmonic Feed-back,
rarely cause trouble except on the short waves or when local
stations are involved. Practically speaking, it is impossible to
obtain a frequency-changer which is completely free from
harmonic generation, so that the only remedy for whistles of
all types, except those due to I.F. Harmonic Feed-back, is an
increase in pre-selection. When the ganging is inaccurate, the
normal degree of pre-selection is greatly lowered, so that in
any case where whistles are found the first thing to do is always
to make sure that the ganging is as perfect as possible.
As a guide to the performance obtainable, it may be said
that in a district 30 miles or so from a local station two signal-
frequency circuits for a no kc/s intermediate frequency, or
one for a 465 kc/s frequency, are adequate. There will be
few, if any, whistles, but there is little factor of safety. At
10 miles from a pair of locals there will be quite a number of
whistles, and to obtain the same performance three signal-
frequency circuits are necessary for the lower intermediate
frequency and two for the higher.
On short waves, second channel interference is almost
inevitable with only one signal-frequency circuit, even with
an intermediate frequency of 465 kc/s. Two are sufficient to
give reasonable protection against it for frequencies up to
some 10-15 Mc/s ; above this three should be regarded as
essential for first class results.
138
SUPERHETERODYNE WHISTLES
In ninety-nine cases out of a hundred, whistles on the
medium waveband will be caused by the local station because
it is so much stronger than other stations. As an example of
the kind of thing to be expected, the signal frequencies at which
whistles may occur due to high-power stations on certain
frequencies are listed in Tables I, II and III for the chief forms
of interference and for intermediate frequencies of i lo kc/s and
465 kc/s. The frequencies of the stations correspond to the
present British station allocations. The superiority of the
higher intermediate frequency in respect of the number of
possible forms of interference is very evident.
An Absorption Wavemeter
The most useful tool in tracing the cause of whistles is a
simple absorption wavemeter covering a band of 100 kc/s to
1,700 kc/s continuously. It should be connected in series with
the aerial lead to the set, the receiver tuned so that the whistle
which it is desired to eliminate appears, and the wavemeter
tuned carefully through its range. One point will always be
found at which both the signal and whistle disappear or are
greatly weakened.
This occurs when the wavetrap is tuned to the wanted
signal and is of no use for present purposes. The aim is to
find a setting for the wavemeter control for which only the
whistle disappears leaving the wanted signal unaffected.
The wavemeter is then tuned to the station responsible
for the whistle, and knowing its frequency and the
frequency of the wanted
signal, the type of inter-
ference can be found.
In some cases, however,
the type of interference
is not very important
for the remedy is the
same for nearly all types
— an increase in pre-
selection.
Now it is not usually
easy to increase the
selectivity of the signal-
frequency circuits ade-
quately, assuming that
the ganging has been
Fig. 71 : A simple wavetrap is often useful in
removing whistles produced by a strong local signal
139
TABLE I
Frequencies upon which Interference may be found from
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141
3rd Harmonic .
WIRELESS SERVICING MANUAL
accurately carried out. As the whistles are usually due to
one or two powerful local stations only, the best remedy
is to include a wavetrap in the aerial circuit and to tune it to
the local station. This reduces the input, as far as the receiver
is concerned, to something little more than the level of a strong
Continental signal and in most cases wipes out nearly all the
whistles. The connections for such a wavetrap are shown in
Fig. 71, and they can easily be made externally to the receiver.
A further advantage of the trap is that by reducing the aerial
input from the local station it greatly lessens the possibility of
distortion when receiving this station. Very strong stations
are sometimes distorted through an I.F. vah^e becoming over-
loaded, for there are limits to the range of control given by
even the best A.V.C. system.
Using Two Traps
In districts where there are two local stations, it will ofteri
be found that the inclusion of a wavetrap removes about half
the whistles only. This is because half of them are due to one
local station and the other half to the other, and the remedy
is to fit two wavetraps, one being tuned to each of the offending
stations.
Trouble may also be experienced from a station working on
the intermediate frequency. With a low intermediate fre-
quency, a high power C.W'. morse transmitter will usually be
the offender, and since high-speed sending is often employed
the audible effect is similar to a steady whistle when a station
is tuned in. Although the interference is usually evident
throughout the tuning range of the set, it is at its worst towards
the upper end of the long waveband.
With an intermediate frequency of the order of 465 kc/s, the
interference is generally from spark transmitters and is at its
worst at the upper end of the medium waveband- and at the
lower end of the long. Here again the remedy is to use a
wavetrap in the aerial circuit tuned to the intermediate fre-
quency.
When a “ large-primary ’’ aerial coil is used an I.F. wavetrap
is almost essential, because the aerial circuit then resonates
close to the intermediate frequency. As the primary is of high
impedance, an acceptor type of wavetrap is more effective than
a rejector. This consists merely of a coil and condenser in
series connected between the aerial and earth terminals as in
Fig. 66 (b). The rejector wavetrap of Fig. 66 (a) or Fig. 71 is
more effective with a small-primary ” aerial coil.
142
SUPERHETERODYNE WHISTLES
Any number of wave-
traps may be used and
connected in series with
one another as shown in
Fig. 72. Here three are
used, one for each of
two local stations and a
third tuned to the inter-
mediate frequency. The
precise values of the
components employed
naturally depend upon
the particular frequen-
cies to which the traps
must be tuned. The
values are readily worked
out from the formulae
given in any reference
book or from Abacs.
The amount of rejection
iinnn f>ffi F«£-72: Three rejector wavetraps can be used as
uepenuS upon tne eni shown here. Two are tuned to local stations and
Ciency of the coils and the third to the intermediate frequency
condensers, but it is
usually possible to obtain good results with small coils and
compression type condensers. Acceptor traps must, of course
be connected in parallel with one another, not in series as with
rejector traps.
Whistles which are due to the feed-back of harmonics of the
intermediate frequency will not respond to wavetrap treatment,
for they are due to stray coupling in the receiver itself. Feed-
back from any I.F. circuit which generates harmonics to any
pre-frequency-changer circuit can cause this trouble, but the
detector is the usual source of the trouble for the signals are
strongest at this point and it necessarily generates harmonics.
The I.F. amplifier should cause no trouble, for it ought to be
linear ; if it is not linear, however, it will produce harmonics
and may cause trouble. The last I .F. valve is usually the danger
point. It may be remarked that this type of interference is
more likely to occur with modern receivers than with older
sets, for it is now the custom to operate the detector at a large
input and this not only makes harmonic production more
likely, but results in a bigger voltage being fed back to the
input through a given amount of coupling.
At first sight it would appear that this trouble is easily
diagnosed, for one would expect whistles to occur only at
143
WIRELESS SERVICING MANUAL
certain definite signal frequencies which are multiples of the
intermediate frequency. This is not so, however, and with a
low intermediate frequency it is possible for there to be a
whistle on practically every medium wave station !
A Band of Frequencies
This effect is brought about by the fact that the I.F. amplifier
passes a band of frequencies and this band width is effectively
multiplied by the degree of harmonic involved. Suppose, for
instance, that the intermediate frequency is no kc/s, and that
high harmonics are troublesome. In a high-quality receiver,
frequencies up to 9 kc/s may be passed, so that the I.F. amplifier
will respond to frequencies between loi kc/s and 119 kc/s.
If it is desired to receive a station on 970 kc/s one would not
at first expect there to be any trouble, for the nearest harmonic
frequency is 990 kc/s, which is 20 kc/s away from the wanted
signal. If the tuning is altered slightly, however, so that the
set is tuned to 968 kc/s, the station still remaining on 970 kc/s,
the intermediate frequency produced by the oscillator beating
with the signal will be 108 kc/s ; the ninth harmonic of this is
972 kc/s, and this will beat with the signal to produce a 2,000
c/s note. Thus although there may be no whistle when the
set is tuned exactly to the station, it will appear with quite a
small degree of mistuning.
Table IV shows the range of frequencies over which this
type of interference may be found with an intermediate fre-
quency of no kc/s. On the long waveband, the band of
202-238 kc/s may contain whistles and, with the exception of
the band of 595-606 kc/s, the whole of the medium waveband.
In some cases, the possible interference bands for each harmonic
overlap. Thus, the 9th harmonic may cause trouble between
909 kc/s and 1,071 kc/s, the loth from 1,010 kc/s to 1,190 kc/s,
and the nth from i,ni kc/s to 1,309 kc/s.
It can be seen that in the band 1,010-1,071 kc/s both 9th
and loth harmonics can cause whistles, while in the range
1,111-1,190 kc/s the loth and nth harmonics may give trouble.
Table V shows the results with a higher intermediate frequency
of 465 kc/s and it can be seen that only the 2nd and 3rd har-
monics can possibly give this type of interference and the
ranges over which whistles may be experienced are confined to
912-948 kc/s and 1,368-1422 kc/s.
In practice, the high harmonics do not usually cause serious
trouble, for they are much weaker than the 2nd and 3rd
harmonics, and if the stray couplings are kept small enough for
144
SUPERHETERODYNE WHISTLES
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I.F. Harmonic Interference
WIRELESS SERVICING MANUAL
only a moderate degree of 2nd and 3rd harmonic interference,
the higher harmonics are likely to be negligible. It is, however,
very difficult completely to eliminate trouble from the 2nd and
3rd harmonics, for in sensitive receivers the merest trace of
coupling between the circuits concerned will lead to prominent
whistles at the appropriate signal frequencies. In bad cases,
actual instability may be set up when the set is tuned to twice
the intermediate frequency.
The remedies are really the same as those necessary in a
straight set suffering from instability, but when testing, it is,
of course, necessary to provide a signal at the appropriate
frequency in order to obtain the whistle. Although a receiver
may be unstable with or without a signal, the worst effects of
harmonic feed-back are only apparent when a signal is present.
In order completely to eliminate the trouble, very thorough
screening of the detector is essential and in some cases of the
I.F. amplifier also. Such screening will be nullified if currents
of the harmonic frequencies are allowed to flow in the con-
necting leads to other circuits. The H.T. supply to the last
I.F. valve must be well decoupled, and sometimes the screen
supply also, while the A.V.C. line must have an initial stage of
filtering placed as closely as possible to the A.V.C. diode and
contained within any detector screening.
The filtering in the A.F. output circuit of the diode detector
must be good, otherwise the harmonic frequency potentials
may be amplified in the A.F. stages and radiated from them.
In some cases, it is necessary to use a tuned filter circuit, as
shown in Fig. 73, tuned to the 2nd harmonic of the intermediate
frequency. The coil L employed in this circuit should be of
suitable inductance to tune to the required frequency with a
capacity C of about 30-50
This form of interference, although objectionable, can
occasionally be turned to good account. When a calibrated
test oscillator is not available the chief difficulty in trimming
an I.F. amplifier is that one has no knowledge of the frequency
to which it is adjusted. It is possible to deduce the frequency,
however, by noting on which stations the whistles caused by
harmonic feed-back occur. By introducing deliberate coupling
between the detector and aerial circuits, as by running the
aerial lead amid the detector wiring, a strong whistle should
be obtained from the 2nd harmonic.
The accuracy is only reasonably high if several whistles can
be found. The set should be tuned accurately to each station
and the pitch of the whistle noted. The station on which the
146
SUPERHETERODYNE WHISTLES
Fig. 73 : In order to eliminite I.F. harmonic feed-back a circuit tuned to
twice the intermediate frequency is sometimes needed in the detector
whistle has its lowest pitch should be selected and the inter-
mediate frequency is then nearly equal to one-half the frequency
of the station. If the whistle has a pitch of about i,ooo c/s,
the error in estimating the intermediate frequency is only
about I kc/s.
147
CHAPTER 15
FREQUENCY-CHANGERS
N early all modem superheterodynes have a frequency-
changer (or mixer) of the multiplicative type, and the
once popular additive type is rarely found in ordinary
apparatus. The multiplicative mixers are those in which the
signal and oscillator voltages are applied to different electrodes
and the alternating anode current depends on the product of the
two control-electrode voltages. In an additive mixer, however,
the signal and oscillator voltages are usually applied to the same
control electrode, and the anode current is then a function of the
sum of the voltages.
Typical examples of multiplicative mixers are heptodes and
hexodes — whether or not combined with triode oscillator
sections — but tetrodes or pentodes with anode or suppressor
injection of the oscillator voltage are also of this type. The
additive mixers are diodes, triodes or any other valve in which
the signal and oscillator voltages are applied to the same control
electrode.
FREQUENCY-CHANGERS
The great practical difficulty with additive mixers lies in the
coupling between the signal and oscillator tuned circuits. This
makes the oscillator frequency dependent on the signal-frequency
tuning, and as a result, ganging becomes quite difficult. The
most successful circuit of this type is the cathode-coupled fre-
quency-changer of Fig. 74. A triode oscillator Vz of conven-
tional design is used and the mixing valve is a tetrode or pentode
Vi adjusted to work as an anode bend detector, cathode-bias
being obtained by Ri. The signal input circuit is LiCi, and
the oscillator voltage is injected into the cathode circuit by
means of the coupling coil L3.
The voltages across LiCi and L3 clearly add between grid
and cathode, and there is at first sight no coupling between the
tuned circuits themselves. In actual fact, however, there is
coupling through the grid-cathode capacity C3 of Vi. Even
with an intermediate frequency as low as no kc/s, the circuit is
quite satisfactory on the medium waveband, and the pulling
between the circuits is small enough to be unimportant.
Pulling is very serious on short waves, however, and the
circuit there becomes almost impracticable. For the best
results Vi should not be of the variable-mu type and it should
not be connected to the A.V.C. system.
A similar circuit to this, but one in which a single valve
acts as a detector-oscillator, is shown in Fig. 75. The signal
149
WIRELESS SERVICING MANUAL
circuit is LiCi and the oscillator L2C2C3. The latter is
fed through C4 from the valve anode and coupled to the cathode
coil L3 which forms the reaction coil. The I.F. transformer
primary is tuned by C4, and the oscillator tuned circuit
actually forms a part of the primary circuit. At the oscillator
frequency, the transformer primary functions as an R.F. choke,
while the I.F. trimmer C4 becomes the feed condenser for the
oscillator circuit.
The circuit is not suitable for short waves, but can be made to
give good results on the medium and long wavebands. C4 is
adjusted in the usual way at the intermediate frequency, for it is
the first I.F. trimmer. The signal and oscillator circuit trimmers
are also adjusted quite normally.
The performance is more dependent upon the kind of valve
used than most circuits, and this is likely to be the greatest
practical difficulty with it to-day. It is only found in sets
of about 1932-1934 manufacture and the valves used are
probably obsolete now. A replacement of different character-
istics will probably demand some juggling with screen and bias
voltages, and even then, the results may not be as good as they
were originally because the grid-cathode capacity will probably
be different.
This type of mixer was also used sometimes in battery sets
and then the circuit usually took the form of Fig. 76. Here
ISO
FREQUENCY-CHANGERS
two coupling coils are used — one in each filament lead —
because there is no separate cathode to the valve. It is
important that these coils be of low D.C. resistance, otherwise the
voltage drop across them will result in the valve being underrun.
In other respects the circuit functions like the mains variety of
Fig. 75. . , . .
The multiplicative mixers fall into two main categories —
heptodes and octodes, with the oscillator voltage applied to an
inner grid and the signal to an outer grid, and pentodes, hexodes,
and heptodes in which the signal is applied to an inner grid and
the oscillator to an outer. These valves usually have a built-in
triode oscillator section, but some have not and require a
separate oscillator ; even those having a built-in oscillator are
sometimes used with a separate one.
The essential difference between the^heptode and the octode
of the first category is that the latter has one more grid. It is
connected internally to the cathode and can be ignored by the
user. It is a suppressor grid and tends to increase the A.C.
resistance of the valve. The essential difference between the
heptode and the octode is thus the same as that between the
screened tetrode and the pentode.
Fig. 77 : This diagram shows the usual connections for a heptode
frequency-changer
151
WIRELESS SERVICING MANUAL
The basic circuit for both valves is shown in Fig. 77.
Numbering the grids in order from the cathode, Gi and G2
form the oscillator electrodes and with the cathode give a triode
oscillator. G2 usually consists of a number of bars mounted
away from the main electron stream, but Gi is a proper grid in
the main stream. This electron stream is modulated by the
oscillator and passes through the inner screen grid G3 on the
outside of which it forms a virtual cathode. The density of
this virtual cathode varies at oscillator frequency since it is
dependent on the instantaneous potential of Gi. Electrons
from it are attracted by the positive outer screen grid G5 and in
their passage to it are controlled by the signal grid G4, which
usually has variable-mu characteristics. Outside G5 the
electrons pass to the anode to form the anode current, the value
of which depends on the instantaneous potential of both Gi
and G4.
There is little special about fault-finding with this type of
mixer. The electrode potentials should be checked in the
usual way and the oscillator amplitude can be measured by
noting the grid current through Ri — again as in any other grid
leak oscillator.
Coupling between the signal and oscillator circuits is by no
means absent, for there are not only small inter-electrode
capacities but there is a negative mutual conductance effect
between Gi and G4. The valve is a great improvement on the
cathode-coupled pentode of Fig. 74 and can be used effectively
on short-waves. Above about 15 Mc/s, however, the coupling
between the circuits is really excessive and although the circuit
can be made to work quite well it becomes a little tricky.
Some octodes include a built-in neutralizing condenser to
reduce coupling effects, and these are a considerable improve-
ment.
The second category of multiplicative mixers is illustrated
by the triode-hexode of Fig. 78. This is a mixing hexode and
a separate triode built into one envelope and having the
cathodes internally connected. The triode grid is also usually
joined internally to the injector grid G3 of the hexode. A
triode-heptode is essentially the same, but has a suppressor
grid between the outer screen grid G4 and the anode. It con-
sequently has a higher A.C. resistance.
In general, the valve is much more suited to short wave
reception than the heptode or octode. Because of the separate
cathode for the oscillator section, the triode part is of higher
mutual conductance and so oscillates more readily. This
152
FREQUENCY-CHANGERS
Fig. 78 : The triode-hexode is now one of the most widely used fre^ueney-
changers, and the circuit is of this general form, but sometimes differs in
minor details
means that the valve will function efficiently at higher frequencies
and also that at frequencies where both are efficient better
frequency stability can be obtained. In addition, the triode-
hexode is superior because the coupling between the signal and
oscillator parts of the valve is lower. In practice, this lower
coupling is achieved only if the amplitude of oscillation in the
triode section is kept below a certain maximum.
These mixers like the triode-hexode, with Gi for the signal
grid, have two disadvantages compared with the heptode type
in which Gi is the oscillator grid. Electron transit times are
greater and at really high frequencies — above about 25 Mc/s —
they have a lower input impedance at the signal grid. They
also take a larger cathode current. This last is unimportant in
a mains set, but is the reason why they are not common in
battery sets.
Instead of a double valve, two separate valves can be used
and at one time an R.F. pentode mixer with separate triode
oscillator was sometimes employed. The suppressor grid was
used for oscillator injection, being frequently connected directly
F 153
WIRELESS SERVICING MANUAL
to the triode grid. The objections to this arrangement are
that a very large oscillator amplitude is needed because the
suppressor grid is usually of too open mesh to give good control
over the electron stream, and the mixer has a low output
resistance and so damps the LF. transformer heavily.
Nowadays, even when two valves are used, the mixing valve
is nearly always a hexode or heptode, the most frequent course
being to use a triode-hexode or triode-heptode and ignore the
triode section. The advantage of using a separate valve for
the oscillator is that it is often possible to obtain better fre-
quency stability.
For a frequency-changer to be satisfactory it is necessary not
only that it be efficient and as free as possible from harmonic
generation, but that the oscillator stability be high. This last
means that the oscillator frequency should be dependent on the
setting of the tuning control only. In practice, this is never
achieved. It is always affected in some degree by its own
electrode potentials, by the electrode potentials of the mixer,
and by temperature. Iffiese are dealt with in detail later, but it
is often easier to reduce them when the mixer and oscillator
valves are entirely separate than when they are combined in one
envelope.
Whatever the form of the mixer, the oscillator itself follows
more or less standard lines and typical circuits are shown in
Fig. 79. Oscillator grid bias is nearly always obtained by using
a grid leak and condenser, as in (aj and (c), but occasionally
cathode-bias (6) is adopted.
When oscillator and mixer are a multiple valve, the tuned-
grid oscillator (c) is nearly always used, but when it is entirely
separate the tuned anode circuit is sometimes adopted. More
often, the cathode- tap Hartley circuit is used and sometimes the
modified Colpitt’s as shown in Fig. 80 (a) and (b) respectively.
The latter oscillates by virtue of the grid-cathode capacity of the
valve and the cathode-earth capacity, which includes the self-
capacity of the choke Ch. This circuit is suitable only for short-
waves.
Whatever kind of oscillatot is used it differs from an amplifier
in one important point way — the anode current has a value
when the valve is oscillating which is different from that when
it is not oscillating. This enables a ready test for an oscillator
to be devised and it is important because the failure of the
oscillator will render any superheterodyne inoperative.
The test is to connect a milliammeter in series with the
anode feed resistance Ri (Figs. 79 and 80), on the H.T. side of it^
154
Fig. 79 ; Broadcast receivers usually have
a tuned-grid oscillator (c), but tuned-
anode circuits (o) and (b) are sometimes
used
and to note the current passing.
The valve must then be placed
in a definitely non-oscillating
condition and it is usually
easiest to do this by short-
circuiting either the grid or
anode coil. If the current is
then found to have the same
value as at first, the valve is
not oscillating, but if a different
value of current is found, then it is quite certain that the valve
oscillates. The direction of the change of current depends upon
the method of biasing.
With grid leak oscillators ( (a) and (c), Fig. 79) the current is
lower when the valve is oscillating than when it is not ; con-
sequently, if all be in order, short-circuiting one of the coils
should cause a rise in anode current. In a typical case, the
anode current of an oscillating valve might be 3 mA, whereas
with the same applied voltages it might be 5 mA when the valve
is not oscillating.
In cases where battery or cathode biasing is used, however,
the current is higher in the oscillating than in the non-oscillating
155
WIRELESS SERVICING MANUAL
condition and the change of current is much larger. A feed
resistance Ri is often essential to limit the current, for if the
possible current consumption is not restricted in some such way,
it can be very heavy indeed ; a small battery valve can easily
take 20 mA or more.
When checking for oscillation, care should be taken to check
at several different points throughout the tuning range, for it
often happens that oscillation is obtained over a portion of the
range only. In general, when the valve is oscillating the anode
current varies somewhat over the tuning range and with a grid
leak oscillator it usually increases at the low frequency end of the
tuning range.
Causes of Non-Oscillation
Non-oscillation may be due to many defects which are in
essence no different from those which may occur in any stage.
Incorrect voltages, a defective valve, a faulty condenser, short-
circuited turns on a coil, can all cause this fault to develop and
can be found by the application of the usual tests. With a
newly-constructed receiver, however, or when alterations to
this portion of a set have been made, the connections to the coils
should always be checked. The correct connections are always
the same and are easily memorised if the tuned and reaction
coils be regarded as a continuous winding which is merely split
at one point for the H.T. and valve cathode connections ; the
two outer ends are then the grid and anode connections. In
other words, if both coils are wound end to end and in the same
direction, the outer ends must be taken to grid and anode. The
same connections hold if one coil be wound over the other, as
can readily be seen if one imagines the coils to be pulled apart
so that they are end to end. With multilayer coils wound in the
same direction, if the outer end of one coil is taken to anode, the
inner end of the other must be joined to grid.
Although it is necessary that the oscillator shall function
smoothly over the waveband, this alone is not sufficient to ensure
satisfactory operation of the frequency-changer. It is necessary
also that the right amplitude of oscillation be applied to the
mixer. When the oscillator valve is quite separate this may be
achieved in the coupling between the two, but with a combined
valve it must be arranged by adjusting the output of the oscillator
itself. With the usual triode-hexode the amplitude can easily
be checked by inserting a microammeter in series with the grid
leak at “ X ** of Fig. 78.
When the valve is not oscillating the grid current will be of
IS6
FREQUENCY-CHANGERS
Fig. 80: The Hartley (a) and Colpitt's {b) oscillators are quite frequently used
in short'Wave equipment
the order of lo ft A. and it will be much higher when it is
oscillating. The change of current in microamperes multiplied
by the value of the grid leak in megohms gives a figure which is
about 83 per cent, of the oscillator R.F. voltage. Thus, if the
change of current is 200 ftA. and the grid leak is 0*05 megohm,
the oscillator apitude is about 12 volts.
The efficiency of the frequency-changer is a function of the
oscillator amplitude. As the latter is increased from zero, the
efficiency rises rapidly, reaches a maximum, and then falls
slowly. The maximum is usually at about 10 volts oscillator
amplitude, but it is high over the range of about 8 volts to 20
volts.
It will often be found that the oscillator amplitude varies
considerably as the set is tuned over a waveband. It often
varies in the ratio of 2 or 3 to i , and it usually falls towards the
low-frequency end of the band. As the maximum of the
oscillator amplitude -efficiency curve is a very flat one, it might
be thought desirable to adjust matters so that the lowest
amplitude is about 8 volts and to let the variations be all above
this on the flat part of the curve.
While it is sometimes permissible to do this on the medium
and long wavebands, it should never be done on short waves.
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WIRELESS SERVICING MANUAL
In the author’s experience the optimum oscillator amplitude is
a maximum which should never be exceeded on short waves.
Excessive oscillator voltage gives rise to coupling between the
signal and oscillator circuits and if the difference between their
resonance frequencies is only a small fraction of the oscillator
frequency, as it is on the short waves, the signal circuit can be
shock-excited by the oscillator under certain tuning conditions.
The effect is a number of very strong spurious signals.
It is brought about by an excessive instantaneous negative
potential on the injector grid of the mixer. The normal
operation of the hexode is that the electron stream from the
cathode is attracted by the positive screen Gz and modulated en
route by the signal grid Gi . Some of the electrons land on G2,
but the majority pass though it to form a virtual cathode outside
G2, the density of this virtual cathode varying in accordance
with the signal applied to Gi. The outer screen G4 attracts
electrons from this virtual cathode and they are modulated in
their passage to it by the oscillator voltage on the injector grid
G3. Some land on G4, but most pass through it and reach
the anode.
Maximum efficiency is obtained when the oscillator ampli-
tude is such that G3 is so negative during most of the oscillator
cycle that the current to G4 and the anode is completely cut-off.
The current then passes in short spurts when the oscillator
voltage swings positively. The oscillator, in fact, acts as an
interrupter.
Now the position of the virtual cathode outside G2 depends
on the potential of G3 as well as on that of G2 and G4. When
the potential of G3 is very negative the virtual cathode is pushed
close to G2. Electrons which come through G2 with a velocity
above the average are turned back and may be so repelled by
the negative grid that they pass through G2 a second time and
find themselves travelling towards Gi on the inside of G2.
They soon turn back towards G2 and either land on it or pass
through it once again to its outside. While inside G2, however,
they induce a charge on Gi and a minute grid current flows.
The resulting current is of a random nature, but as it can only
occur when the injector grid potential is more negative than a
certain value, it is definitely related to the oscillator frequency.
The nearest simple description is perhaps to call it a random
current interrupted at oscillator frequency.
Even if the oscillator amplitude is such that electrons do not
actually return through G2, the change in position of the virtual
cathode.outside G2 is sufficient to influence Gi to some extent,
because the screening effect of G2 is not perfect.
158
FREQUENCY-CHANGERS
The net result of all this is to give some coupling between
the signal and oscillator circuits and to set up some oscillator
voltage on Gi The oscillator voltage on Gi increases as the
ratio of the signal and oscillator frequencies gets nearer to
unity — that is, as the frequency increases — and the first result
is to produce grid current at low signal grid biases. It is, there-
fore, common practice to use more grid bias on Gi on short
waves than would be necessary on the me«iium waveband. The
second result is to produce spurious signals which are often
suggestive of instability of the R.F. amplifier.
To avoid these effects it is most important to limit the
amplitude of oscillation to a maximum which is usually about
lo volts, and to let any variations in the amplitude occur below
this figure. Variations of the order of 3-1 can no longer be
tolerated, however, for the efficiency would suffer severely.
Constant Oscillator Output
It is necessary, therefore, to obtain a more even output from
the oscillator and there are several expedients which can be
adopted. In general, the oscillator output is at a maximum at
or near the high-frequency end of the tuning range because,
with variable condenser tuning, the dynamic resistance of the
tuned circuit increases with frequency. One simple way of
reducing the variations of output is to shunt the tuned circuit
with a resistance. If this is equal to the dynamic resistance at
the low-frequency end of the band, the variation of total dynamic
resistance over the band must be less than 2:1. The output
at all points is reduced, but can be brought back to normal by
increasing the reaction coupling and/or the H.T. voltage.
While sometimes useful, this scheme is not a good one
because the tuned circuit is more heavily damped and the
frequency stability is consequently reduced. Better alternatives
are to connect resistances in series with the grid or anode leads
of the valve. In conjunction with the valve capacities they form
attenuators giving more attenuation at high frequencies than
at low, and so even up the output. The values depend on all
the other circuit constants and must be found by trial, but
50 ohms to 1,000 ohms is usual. Unfortunately, even these
resistances reduce frequency stability, for not only do they damp
the tuned circuit, especially at the high-frequency end of the
band, but they cause a phase change in the oscillator circuit.
Apart from parasitic oscillation and spurious signals, which
must at all costs be avoided, the most important thing in a
159
WIRELESS SERVICING MANUAL
frequency-changer is the frequency stability of the oscillator. If
the set is not to require continual re-tuning, it is essential that
any incidental changes of oscillator frequency be small compared
with the band-width of the I.F. amplifier. Thus suppose
that the band-width is lo kc/s. When a signal is tuned in it
will be centred on the pass-band and both sidebands up to
5 kc/s will be passed. Now if the oscillator drifts by i kc/s the
carrier will be i kc/s off the centre of the pass-band and side-
bands up to 4 kc/s on one side and 6 kc/s on the other will be
passed. Double-sideband reception up to 4 kc/s only will
then be possible, with an approach to single-sideband reception
up to 6 kc/s. There will be some change of quality, but in this
case not very much.
Suppose, however, that the drift reaches 5 kc/s. Double-
sideband reception will be confined to the lower frequencies,
and there will be single-sideband reception up to 10 kc/s.
There will then be considerable amplitude distortion because
the linear detector required for double-sideband reception is
very far from right for single sideband. If the drift is any
greater, the carrier will fall outside the pass-band and its
amplitude will fall very rapidly ; it may easily become weaker
than the sidebands. In general, if distortion is to be small the
frequency drift must not exceed some lo per cent, of the band-
width. On short waves, where the signal is probably already
distorted by selective fading, probably 20 per cent, drift can be
tolerated.
Drift is especially important in push-button sets, because
unless A.F.C. is fitted, the user has no way of correcting for it.
It must then be kept always below 10 per cent, of the band-
width.
Oscillator drift arises from many different causes, but there
are always small changes in the values of the circuit “ constants.^’
The oscillator inductance and capacity are both affected by
temperature, while the mutual conductance and A.C. resistance
of the valve are affected by the voltages applied to the electrodes.
Any factor which alters the effective resistance or reactance of
the oscillator tuned circuit affects the frequency.
The degree of stability required is very high indeed on short
waves and is by no means easy to achieve. On the medium
waveband, i kc/s drift corresponds to a stability varying from
I part in 1,000 at the low frequency end to i part in 2,000 at the
high frequency end. At 30 Mc/s (10 metres), however, it is
z part in 30,000.
The frequency variations are of two kinds — slow and rapid.
160
FREQUENCY^CHANGERS
The slow changes are usually caused by temperature effects.
When a set is first switched on the temperature of all its parts
is at the ambient, i.e., the temperature of the room. Heat is
generated in the cathode of the oscillator and as a result of
expansion the inter-electrode capacities change. The major
part of this change occurs before the valve is properly operative,
and is unimportant, but small changes may still occur for some
minutes afterwards. The effect on frequency depends largely
upon the fraction of the total tuning capacity which the changes
represent. To keep the effect small the tuning capacity should
be as large as possible.
In addition to the oscillator, all the valves in the set produce
heat, and the mains transformer, speaker field, and resistances
contribute their quota. A typical mains set consumes 6o to
loo watts and all but the minute proportion radiated as sound
by the loud speaker is dissipated as heat within the cabinet. It
is, therefore, inevitable that the temperature inside the cabinet
should rise above the ambient.
Ventilation
The extent of the rise depends chiefly upon the arrangements
for ventilation and with good design it can be kept small. With
good design, too, the set will attain its final temperature quickly.
Unless temperature compensation is adopted, the oscillator
frequency will drift while the temperature is changing. In
some of the older sets it is not uncommon for the oscillator to
drift 5 kc/s or so in the first 15 minutes after switching on, even
on the medium waveband. In such sets it may take an hour
or so for real stability to be reached.
The layout of the parts and the ventilation have an enormous
effect upon the stability. This is easily seen by considering an
extreme case. Suppose the set is completely sealed so that there
is no ventilation at all, and suppose also that the case is perfectly
lagged so that there is no loss of heat through its walls. Heat
is continually being generated inside the case, and there is no
loss of heat ; therefore, the temperature rises continually.
Frequency stability is then never achieved.
If the case is not perfectly lagged, the interior will eventually
reach a temperature such that the loss of heat through its walls
equals the rate of internal heat generation. A stable condition
then exists and drift will cease. If ventilation is provided,
however, heat will be lost more easily, and the stable condition
will be reached at a lower temperature and more quickly. The
161
WIRELESS SERVICING MANUAL
drift will be reduced in magnitude and reach its maximum
value more quickly.
The main principles in reducing thermal drift are to get the
heat necessarily produced in the frequency-changer away from
it as quickly as possible and to prevent heat generated in other
parts of the set from reaching it. The first thing is to mount
especially hot parts like output valves, rectifiers, speaker fields
(and sometimes mains transformers !) and hot resistances as
far as possible from the frequency-changer circuits. The
second thing is to give these hot parts plenty of ventilation so
that the heat is taken right outside the cabinet. This is best
done by providing holes in the case immediately above and
below the hot parts and adequately large holes in the chassis.
For an output valve of 10-15 watts anode dissipation a hole
ij in. diameter immediately above it and another below it are
adequate if the chassis has holes of the same total effective area
around the valveholder. The holes in the case must usually be
covered with gauze, so that the effective area is reduced, and
becomes perhaps only i sq. in. Eight holes of 0*4 in. diameter
spaced around the valveholder will provide an unimpeded path
for the air flow.
Never forget the chassis holes. It is no use making a large
outlet at the top of the set and a large inlet at the bottom, if the
chassis between the two obstructs the free flow of air between
them.
If the construction is such that it is not possible to space the
hot parts widely from the frequency-changer, then having
provided ventilation as described above, it will pay to erect a
heat screen around the hot parts. This should take the form of
a vertical metal screen fitting tightly between the chassis and
the case. As the hot parts are usually above the chassis, it is
generally sufficient for the heat screen to be above the chassis
only, but if there is anything hot underneath, then a screen will
be needed there too. The screen should be of bright metal,
preferably polished, on the side facing the heat. It is some-
times advantageous to lag the other side with heat insulating
material, but is often unnecessary.
The frequency-changer itself should be laid out so that its
parts are in as cool a place as possible. It is usually best to
mount the gang condenser above the chassis and alongside the
valve, with the oscillator coil and all trimmers and switches
under the chassis beneath the gang condenser. Any resistances
generating an appreciable amount of heat, such as H.T. dropping
resistances and Screen-feed potentiometers, should be mounted
162
FREQUENCY-CHANGERS
above the chassis and well away from the gang condenser.
The valve will generally need a screen for electrical reasons
and the opportunity should be taken of turning it into a chimney
for ventilation. To this end it should be of such height that
it almost touches the top of the cabinet, which should have a
hole cut in it immediately above. The top of the valve screen
must not be solid, as it usually is, but must be covered with
gauze or have a large number of holes drilled in it. The chassis
around the valveholder and inside the screen must be drilled
with a large number of holes to allow the free entry of air. The
inside of the can should be of bright metal.
This screen not only provides the electrical screening neces-
sary, but prevents the valve heat from reaching the gang con-
denser and keeps the valve temperature itself at a minimum by
the cooling effect of the draught up the chimney.
The main practical difficulty in providing a set with adequate
ventilation lies in disguising the appearance of the outlet holes
on the top of the cabinet. Some ingenuity is often needed,
but if sufficient trouble is taken it can be done, and it is well
worth while, for in addition to reducing frequency drift the
general temperature is reduced and components are likely to
have longer lives.
A radio-gramophone is particularly difficult to ventilate
because the motor and turntable usually prevent one from
placing the outlet holes above the set. The best thing to do then
is to place the receiver right at the bottom of the cabinet and to
fit a deflector plate above it to guide the hot air to an outlet at the
top of the back of the cabinet. Alternatively, give the set itself
a closed case with ventilation holes at top and bottom, and pro-
vide “ chimneys ” from the top holes to outlets at the top-back
of the cabinet proper.
Even when all has been done that can be done in the way of
ventilation there will still be some frequency drift with tempera-
ture. Even if there were no internal heating, there would still
be changes in the ambient to consider. There is a considerable
difference between an unheated room on a cold day and that
same room an hour after lighting a fire !
Whether or not the drift so caused is important depends on
the band -width and the signal frequency. On the medium
waveband and with a moderately wide pass-band, it is not very
important, except in push-button sets where the user cannot
re-tune if there is excessive drift. On short waves with a narrow
pass-band it can be very important.
Temperature compensation is often adopted in order to pre-
vent changes in the ambient temperature and unavoidable
163
WIRELESS SERVICING MANUAL
internal heating from causing frequency drift. Normal com-
ponents in the oscillator circuit have positive temperature
coefficients ; that is, their capacities and inductances increase
with a rise of temperature. It is, however, possible to obtain
condensers with a special dielectric having a negative tempera-
ture coefficient.
By including such condensers in the oscillator circuit it is
possible so to balance the positive and negative temperature co-
efficients that the net coefficient for the circuit as a whole is
zero. It is usually possible to do this only at one frequency,
but by making this frequency towards the middle of the wave-
band a considerable reduction of drift over the whole band can
be secured. In push-button sets of the type in which switching
is used to select a different oscillator circuit for each station,
individual compensation is possible on each station.
The temperature compensating condenser is sometimes made
adjustable, but its adjustment is apt to be a rather lengthy
business. After each alteration to it it is necessary to put the
set through a complete cycle of temperature change in order ^o
determine its effect. Such a cycle can hardly take less than an
hour, and may be much more.
In general, if the set has been compensated correctly in the
first place subsequent adjustments are needed only if a valve or
component in the oscillator circuit has been replaced by one of
different type. The compensation, therefore, should normally
be left alone unless a drift test shows that its needs attention.
Such a test should always be made, for excessive drift is a
particularly annoying defect.
In addition to the slow changes of frequency produced by
thermal effects, short period changes are often found. These
are usually caused by variations in the voltages applied to the
oscillator and mixer valves. For maximum stability the
oscillator valve should be of high mutual conductance and very
loosely coupled to the tuned circuit which should itself be of
high Q. The amplitude of oscillation should also be kept as
small as possible. As pointed out earlier, therefore, it is im-
portant that any methods adopted to secure an even oscillator
output over the tuning range should damp the tuned circuit as
little as possible.
Sometimes quite elaborate arrangements are made to achieve
high stability, but the details of design are usually by no means
obvious. With some circuits, for instance, it may well happen
that an increase of voltage on one electrode of the valve may
cause a reduction of frequency while an increase on another may
164
FREQUENCY-CHANGERS
Fig. 81 : A simple neon-stabiliser for the oscillator H.T. supply often con-
siderably improves the frequency stability
produce an increase. By proper choice of the feed resistances
it can then be arranged that the change of voltage on one elec-
trode is corrected by the change on the other, and a variation in
the H.T. supply voltage produces no change of frequency. In
such a case the circuit diagram gives no indication at all that the
set has such compensation.
In general, compensation of this nature is rarely more than
partial in broadcast receivers and variations of the H.T. voltage
do affect the frequency to some extent. It is, of course, quite
easy to stabilize the H.T. supply to the oscillator, for the voltage
and current required by the valve are usually quite small. A
neon tube can be used, as shown in Fig. 8i. Special neon
stabilisers are usually made to work at about lOo volts and are
available for a number of different current ratings. One for
some lo mA. is generally suitable, and in such a case one would
choose Ri to drop the H.T. line voltage to loo volts at lo mA
and Rz to drop from loo volts to the required oscillator voltage
at the oscillator current.
As an example, suppose that the H.T. line is at 250 volts,
and that the oscillator requires 5 mA. at 60 volts. With the
above neon, Ri must drop 150 volts at lomA., so it must be
15.000 ohms ; Ra must drop 40 volts at 5 mA. and must be
8.000 ohms.
The tube functions because it has a very low differential
resistance with a relatively high D.C. resistance. If the H.T.
voltage rises the current through the neon increases so much
165
WIRELESS SERVICING MANUAL
that the voltage drop across Ri increases very nearly as much as
the increase of H.T. voltage ; the voltage across the neon,
therefore, rises by a very small amount only.
The degree of stabilisation obtainable depends on the neon
tube, but also on the value of Ri, and it increases with Ri.
Ordinary neon lamps used for lighting can be employed if they
can be obtained without the resistance which is usually fitted
into the base. The working voltage must be found by trial,
however, and it is sometimes as high as 170 volts.
Stabilisation of the oscillator anode voltage alone does not
necessarily improve the frequency stability of a complete
frequency-changer, however, for voltage changes on the mixer
electrodes may affect the frequency in the opposite way to
changes on the oscillator itself. Strictly, therefore, stabilisation
of the whole H.T. supply to the mixer is desirable, but it is not
always easy because the mixer takes a total current of 10-20 mA.
at up to 250 volts. Generally speaking it is impracticable unless
the H.T. line is at least 350 volts.
At the present time it is most unusual to find H.T. stabilisers
in broadcast receivers, and the designer selects values for the
various feed resistances on the frequency-changer to give the
maximum stability. It is, therefore, unwise to change resistance
values or the type of valve used, since although the performance
in other respects may be unaltered or even improved, such
changes may well affect the frequency stability.
Effect of A.V.C.
Some of the most serious effects on frequency stability arise
through A.V.C. and their cause is often unrecognised. The
oscillator frequency can be affected bv the A.V.C. voltage both
directly and indirectly. The direct affect arises when A.V.C.
bias is applied to the signal grid of the mixer for it alters the
space charge distribution in the valve, and hence the input
capacity of the injector grid It also changes the currents
drawn by the various electrodes, and so the voltages applied
to them.
The effect is usually less with mixers having an inner grid
for the signal, such as the triode-hexode, than with those like
the heptode in which the signal is applied to an outer grid.
Nevertheless, it is the general practice wherever possible to
avoid applying A.V.C. bias to the mixer. It is usually only in
sets having an R.F. stage that one can avoid it, however, and in
the smaller sets, with a mixer and one I.F. stage, A.V.C. must
166
FREQUENCY-CHANGERS
be applied to the frequency-changer if it is to be of any real use.
The indirect effect of A.V.C. on the oscillator frequency
occurs even when A.V.C. is not applied to the mixer, and it
occurs because the regulation of the H.T. supply is not perfect.
Changes of A.V.C. bias vary the current drawn by the controlled
valves, and so the load on the H.T. supply. The H.T. voltage
thus varies by an amount depending on its regulation and the
H.T. supply to the mixer varies, so affecting the oscillator
frequency.
In the absence of a stabilised supply for the mixer, all one can
do is to see that the mixer is fed from a point on the H.T. supply
system a^ which the regulation is as good as possible. To this
end, the oscillator is sometimes fed through resistance-capacity
smoothing circuits directly from the output of the H.T.
rectifier. Only the rectifier itself and the mains transformer
are then common to all valves and the oscillator anode voltage is
much less affected.
The effect of A.V.C. on instability often passes unnoticed
because it is troublesome only on a fading signal. On such a
signal the varying oscillator frequency results in distortion
which is very like that caused by selective fading, and the cause
is usually attributed to such fading.
In order to check a set for this effect a signal from a test
oscillator should be applied to the input and its output varied
over a wide range while continually checking the tuning. If
the tuning varies with input, then the A.V.C. voltage is affecting
the oscillator stability.
It is very important that the test oscillator be one in which
the output control has no effect on frequency, for otherwise the
test is completely useless. This rules out many otherwise
excellent test oscillators, for in the cheaper types it is not un-
common to find that the output control affects the test oscillator
frequency more than A.V.C. affects the set oscillator.
Lest it should be thought that too black a picture has been
painted of the difficulties of achieving good frequency stability,
it may be as well to add that although the effects described are
always present their magnitudes are often unimportant at the
lower frequencies. On the medium and long wavebands it
is quite rare for frequency stability to be troublesome if the
ventilation is adequate. Temperature compensation is adopted
only in push-button sets in most cases. As the frequency
increases so does it become more difficult to achieve adequate
frequency stability, and above some 15 Mc/s all the efects
become quite important.
167
W/RELESS SERVICING MANUAL
168
Fig. 82 : The delayed diode A.V.C. system is the most widely used. Three controlled stages are shown, but less bias is applied to the
CHAPTER 16
AUTOMATIC VOLUME CONTROL SYSTEMS
M ost modem receivers include some form of automatic
volume control, and although defects are quite rare,
when they do occur they are usually rather obscure.
In some cases, a defect will result in a loss of sensitivity, while
in others severe distortion may arise. All A.V.C. systems in
common use act on the same basic principle that the increase
in detector input brought about by an increase in signal strength
results in the application of a greater negative bias to certain
valves. This reduces their amplification, with the result that
the detector input rises less than it would do if the control
were absent.
Although the ultimate aim of an A.V.C. system is the main-
tenance of a constant detector input irrespective of the signal
strength it is obvious that this can never be achieved when the
A.V.C. bias voltage is derived from the detector input, for if
the detector input were always the same, the A.V.C. bias and
hence the pre-detector amplification, would be also. In
practice, therefore, the detector input does rise with increasing
signal strength, but in a well-designed system it may rise by
only 6 db. or so for an increase in aerial input of as much as
70 db, or more.
The most widely used method of obtaining A.V.C. is the
delayed diode system, the usual connections for which are
shown in Fig. 82. The precise arrangements employed differ
from set to set, but all using this method of obtaining A.V.C.
are basically the same. The output valve shown is a duo-
diode-triode, in some sets a simple duo-diode is used, in others
a duo-diode-output pentode, while the diodes are sometimes
incorporated in the last I.F. valve. This is not important,
because it is only the diodes that really matter as far as A.V.C.
is concerned.
It will be seen that one diode is fed from the last I.F. trans-
former secondary and that its load resistance is returned
directly to the cathode. This is the detector, and the A.F.
potentials are taken off to the A.F. amplifier, in this case the
triode section of the valve, from the slider of the volume
control Rg. The lower end of this resistance is taken to a
point negative with respect to the cathode in order to obtain
the correct negative grid bias.
A.V.C. id obtained from the other diode which is fed from
169
WIRELESS SERVICING MANUAL
the primary of the I.F. transformer through C5, *which is
usually of about 0*0002 /uF capacity. Its load resistance com-
prises R5 and R6 in series and is returned to a point considerably
negative with respect to the cathode, but which is also the
correct point for returning the grid leads of the controlled
valves. Considering Fig. 82 in the absence of a signal, it will
be seen that the grid of each controlled valve is taken to the
earth line through a series of resistances.
Grid Potential
As there is no current flowing through these resistances, the
steady potential of each grid is the same as that of the earth
line, and each valve has a negative grid bias with respect to its
own cathode of a value depending upon the voltage drop across
its own cathode bias resistance. Furthermore, while the triode
section of the last valve is biased by its biasing resistance R7,
the A.V.C. diode has a negative bias which is the sum of the
voltage drops along R7 and R8 due to the anode current of
the triode.
In some cases, R8 is omitted, and the A.V.C. diode bias is
then equal to the drop across R7 alone. This A.V.C. diode
bias is known as the A.V.C. delay voltage.
Now when a signal is tuned in nothing happens to disturb
this voltage and current distribution as long as the peak I.F.
input to the A.V.C. diode is less than the delay voltage, and
the receiver functions just as if it were not fitted with A.V.C.
When the input exceeds the delay voltage, however, rectification
occurs in the A.V.C. diode circuit and a current flows through
the diode load resistance (R5, R6) in such a direction that the
A.V.C. diode anode acquires a potential negative with respect
to the earth line.
This potential is communicated to the grids of the two first
valves through the resistances R4, R2 and Ri as increased grid
bias, so lowering their amplification. Sometimes the grid
return circuit of the last I.F. valve is also connected to this
point, sometimes it is connected directly to the earth line and
is not controlled. More often it is taken to a tapping on the
A.V.C. diode load resistance as shown in Fig. 82 ; the A.V.C.
bias applied to this valve is then less than the full voltage across
R5 and R6 and is equal to the full voltage multiplied by
R6/(R5 4 - R6).
This is done in order to prevent the last I.F. valve from
being overloaded on a strong signal, for the maximum un-
distorted output obtainable from a screen-grid or R.F. pentode
170
AUTOMATIC VOLUME CONTROL SYSTEMS
valve ia usually much less at a high grid bias than at a low.
Where maximum immunity from such distortion is desired,
the last valve is not controlled at all.
Defects in A.V.C. Systems
In some cases the A.V.C. diode is fed, not from the primary
of the I.F. transformer as in Fig. 82, but from the secondary,
and the diode feed condenser C5 is then snapped between the
two diode anodes. This arrangement is not much used nowa-
days, for the risk of distortion and sideband screech occurring
when the set is mistuned from a station is greater.
Now it Is clear that with such a simple A.V.C. system there
is really very little that can go wrong, and it is for this reason
that defects are rare. Since the delay voltage depends on the
anode current of one of the valves in most cases — in Fig. 82
the triode section of the duo-diode-triode — a loss of emission
in this valve will result in a lower delay voltage. A.V.C. will
then cease to hold stations of different strengths at nearly the
same volume and signals will be generally weaker.
The effects are, however, by no means so noticeable as the
distortion arising in the A.F. amplifier through the defective
valve, and the fault is more likely to be found while tracing
distortion than while testing the A.V.C. circuit. In any case,
the delay voltage will naturally be one of the first things to be
checked in common with the other voltages.
The resistances associated with the A.V.C. circuit are usually
of high value ; Ri to R6 are often 0-5 megohm to 2 megohms
each, so that a slight leak in one of the by-pass condensers or
defective insulation of the A.V.C. line or grid circuits may
materially reduce the A.V.C. bias applied to the valves. Thus,
if R4 is I megohm and C4 develops an internal leak of
5 megohms, the bias actually applied to the valves controlled
from this line will be five-sixths of the full voltage across the
load resistance.
This would appreciably reduce the effectiveness of A.V.C.,
but would not in many cases be very serious. If the leak in
C4 were of 0*5 megohm, however, only one-third of the total
voltage would be available and the performance would be
greatly upset. Similarly, an internal leak in Ci and C2 would
affect the results in like manner, as would also poor insulation
in the grid circuits of any of the valves.
Another danger point is C5. Very high insulation is needed
here, for one side is in connection with positive H.T. A small
leak will slightly reduce the delay voltage actually applied to
171
WIRELESS SERVICING MANUAL
the A.V.C. diode and will not in itself be serious. It will
result, however, in a positive bias being applied to the A.V.C.
line in the absence of a signal, and this may offset the initial
negative bias of the controlled valves sufficiently to promote
instability. A heavy leak in C5 results in the A.V.C. diode
being continuously conductive, and the A.V.C. line takes up a
slightly positive potential.
• It is wisest to test for leaky condensers while they are in the
set, as equipment for measuring resistances of over i megohm
is not usu^ly available. The feed-condenser C5 is readily
tested if a milliammeter be connected in the anode circuit, or a
voltmeter be joined across the bias resistance, of one of the
controlled valves. With no signal, disconnect C5 from the
anode of the preceding valve at the point X. If the anode
current or grid bias of the controlled valve falls with C5 dis-
connected, the condenser is definitely leaking and should be
replaced.
In the case of the condensers in the A.V.C. line, the test is
best carried out with a signal which is most conveniently
obtained from a test oscillator. This can be set at any desired
frequency, the set tuned to it, and the output set at a figure
which gives a good deflection on a meter connected to any of
the controlled valves. Then disconnect C4 at Y ; there should
be no change in the meter reading. If the current falls, how-
ever, indicating an increase of A.V.C. bias, C4 is leaking.
The other condensers, Ci, Ca and C3 can be tested in a
similar way, but here it is necessary for the indicating meter to
be connected to the particular valve concerned, and the test
signal ??iust be injected at a point following this valve. If this
is not done, the removal of the condenser will affect the R.F.
input to the detector and hence the A.V.C. voltage.
Although at one time nearly universal, this form of delayed
diode A.V.C. is gradually falling into disuse because it causes
a degree of amplitude distortion which can be quite considerable
under certain conditions of signal strength and modulation
depth. This occurs because the input resistance of a diode
with negative bias on its anode varies with the input signal.
With no signal it is infinite, and with a very large signal it
becomes, in the limit, one-half of the value of the load resistance.
There is, moreover, an abrupt transition from an infinite to a
finite resistance when the signal changes from a value just
below to a value just above the negative anode bias.
Now a modulated signal is one which is continually varying
in amplitude and as it does so the input resistance of the A.V.C.
172
AUTOMATIC VOLUME CONTROL SYSTEMS
diode varies also, and jumps to infinity whenever the modulation
makes the signal fall below the delay voltage. This input
resistance damps the tuned circuit, and as it varies, the damping
varies also. The gain of the I.F. valve consequently changes
in sympathy. If the gain thus varies during the modulation
cycle the modulation envelope is distorted, and the A.F. output
from the detector is also distorted.
The amount of distortion depends on the relative circuit
values, and it can be minimised by using high values for R5,
R6, R3 and R4 in Fig. 82. together with an I.F. transformer
for the last stage which has coils of low dynamic resistance.
These conditions were often met in the older sets, but not in
more modern designs. The use of iron-core coils has resulted
in higher values of dynamic resistance, and at the same time,
the resistances in the A.V.C. circuit have to be lower. This
is because the maximum D.C. resistance in the grid circuit of
most modern valves is limited to 3 megohms. If a resistance
is common to more than one valve, its effective value per valve
is its actual resistance multiplied by the number of valves.
If this rule is adhered to, the resistances sometimes become
surprisingly low and the distortion correspondingly high.
Alternative circuits are coming into use, therefore, and one of
the most satisfactory is shown in Fig. 83. In this Vi is the
last I.F. amplifier and V2 is a diode detector. The detector
circuit is conventional with a filter comprising Ci, C2 and Ri.
The D.C. load is R2 which is also the A.F. volume control,
and the A.F. output is taken off through C3 and R3. The
detector circuit as a whole is biased positively with respect to
the earth line by R5 and R6, but maintained at earth potential
to alternating currents by C5, which should be of large capacity.
In general the diode circuit will be about 10 volts positive.
A.V.C. is taken off the diode load through R4, and C4 is the
usual filter condenser. Delay is provided by the positive bias
on the detector circuit in conjunction with the diode V3. In
the absence of a signal, the positive bias is applied to V3 through
R2 and R4. This diode is conductive, therefore, and in effect
short-circuits the A.V.C. and earth lines. As the signal gets
stronger the detector output increases, and as it appears across
R2, it is effectively in series with the positive bias voltage as
far as V3 is concerned. When it exceeds the positive bias the
anode of V3 becomes negative with respect to its cathode and
the valve becomes non-conductive and has no further action.
The voltage across R2, less the positive bias, is then applied
to the A.V.C. line.
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WIRELESS SERVICING MANUAL
As in the absence of a signal the controlled valves have their
grids returned to earth only through their filter resistances and
the diode V3, and as this diode is of low resistance, the limitation
on the maximum permissible resistance values is somewhat less
stringent. It is usually satisfactory to take the limit as
2 megohms per valve, for resistances between the grid of a
controlled valve and the anode of V3.
The circuit is not quite free from a similar form of amplitude
distortion to that found with the older arrangement. When
the signal is less than the delay voltage, the D.C. load resistance
of the detector diode V2 is Ri R2R4/(R2 R4), whereas
with a signal greater than the delay voltage it becomes Ri -f R2.
The change of resistance reflects on to the input tuned circuit
and so alters the I.F. stage gain.
The distortion is usually much less than with the older
circuit, however, and is minimized by making R2 as small and
R4 as large as possible. In practice, the two diodes are usually
AUTOMATIC VOLUME CONTROL SYSTEMS
the two halves of a duo-diode, but the valve must be of the
type, such as the 6H6, having separate cathodes.
D.C Amplified A.V.C.
D.C. amplified A.V.C. is used in some receivers because it
gives much greater control than the more simple delayed diode
system. It has the drawback, however, of being much more
particular about its precise operation conditions. The basic
circuit is shown in Fig. 84 and it will be seen that the circuit
of the detector diode is normal, the load resistance Rz also
acting as a manual volume control. The triode section of the
valve, however, has no fixed negative bias applied to it, but its
grid is connected through Ri , which is usually about 2 megohms,
to the detector load resistance Rz. When a signal is tuned in,
a steady voltage nearly proportional to the I.F. input to the
detector appears across Rz and is consequently applied through
175
WIRELESS SERVICING MANUAL
Ri to the triode grid as negative grid bias. The anode current
of the triode consequently falls with increasing signal input.
In the absence of a signal, the circuit constants are chosen so
that the cathode is positive with respect to the earth line and
since the A.V.C. diode anode is returned to the earth line it is
negative with respect to the cathode by this amount — the delay
voltage — and is non-conductive. The A.V.C. line is thus at
the same potential as the earth line.
When a signal is tuned in, the triode anode current falls as
already explained, and the voltage drop across the cathode
resistance R3 falls also, so that the potential of the cathode
comes nearer to that of the earth line. Nothing else happens
until the signal is strong enough to bring the cathode potential
below that of the earth line ; that is, until the voltage drop
across R3 is less than the voltage between negative H.T. and
the earth line.
The A.V.C. diode anode then becomes positive with respect
to the cathode and this diode conducts. The A.V.C. line then
takes up very nearly the same potential as the cathode and with
increasing signal strength follows the cathode potential, becom-
ing more and more negative with respect to the earth line.
The control obtained with a circuit of this nature can be
very effective indeed. The detector, of course, must work at
quite a small input of i or 2 volts only, otherwise the triode
section of the valve, which works as an A.F. amplifier, in
addition to being a D.C. amplifier for A.V.C. purposes, will be
badly overloaded. Apart from leakages in the decoupling
condensers which have the same effect as with Delayed Diode
A.V.C., the chief troubles are likely to arise from a defective
duo-diode-triode .
The A.V.C. Delay Voltage
It can be seen that quite small changes in values will affect
the operation, for the A.V.C. delay is large and depends entirely
on the anode current of the valve. A common value for R3
is 20,000 ohms and the anode current may be 4 mA ; the
cathode potential is then 80 volts above negative H.T. The
voltage between negative H.T. and the earth line may be
50 volts, 80 that the delay is 30 volts. Suppose now that the
valve only passes 3*5 mA without a signal, the delay falls to
70 — 50 = 20 volts ; the performance of the receiver would
obviously be affected.
A serious loss of emission would render the set practically
useless. If the anode current, in this case, fell to 2 mA through
176
AUTOMATIC VOLUME CONTROL SYSTEMS
such a cause, the cathode potential would be only 40 volts
above negative H.T. and would be 10 volts negative with
respect to the earth line. All the controlled valves would then
receive an initial bias of nearly 10 volts which would increase
still more on tuning in a signal.
I'he set would naturally prove very insensitive and would
probably fail to receive anything but a local station. When
testing a set fitted with this form of A.V.C., therefore, it is
particularly important to investigate the operating conditions of
the duo-diode-triode which provides A.V.C.
A few receivers employ I.F. amplified A.V.C. In this
system, a conventional delayed diode A.V.C. circuit is employed
but preceded by a stage of I.F. amplification which operates to
feed the A.V.C. diode only and is not in the chain leading to
the detector. The usual connections are illustrated in Fig. 61
and the possible defects in practice are those of the delayed
diode system with the addition of those likely to occur in any
stage of amplification.
In some receivers, metal rectifiers are used in place of valve
diodes. The circuits are essentially the same, and fault finding
consequently follows the same laws. It should be remembered,
however, that the internal resistance of a metal rectifier in the
“ non-conductive direction is by no means infinite and in
many cases this has the effect of reducing the initial bias of the
controlled valves below the voltage set up across the bias
resistance.
In sets using such rectifiers one should be prepared to find
that the apparent initial bias, as measured across the bias
resistance, is higher than is usual for the type of valve used,
for the designer will have raised the value of the initial bias
resistance in order to compensate for the leakage current of
the rectifier.
Q.A.V.C.
Turning to Q. A.V.C. systems, these are so varied that it is
hardly possible to deal in any detail with the defects which
may be encountered. Basically, the muting circuits are simple,
but many are highly complicated in their detail and as there is
no approach to standardization it is usually necessary to analyse
each case on its merits. In practically all cases muting is
obtained by arranging for a valve to apply a large bias to the
last 1 .F. valve, the detector, or the first A.F. valve in the absence
of a carrier, so that the stage is rendered inoperative.
The I.F. valve is not often silenced, for although it is from
177
WIRELESS SERVICING MANUAL
many points of view the ideal arrangement, it demands the
duplication of this stage and so it is more expensive.
One of the simplest systems is shown in Fig. 85 and if the
operation of this be fully understood little difficulty will be
found with other arrangements of similar type but of different
detail. The last I.F. valve is of the R.F. pentode-duo-diode
type and provides delayed diode A.V.C. in the usual way, the
A.V.C. bias available after the usual filter (R2 Ci) being applied
to the grid of the muting valve V as well as to the early stages
of the receiver.
The detector circuit is normal save for the connection to the
anode of V and for the return of the cathode circuit to the
potentiometer across the H.T. supply instead of to the earth
line. It is clear from the drawing that the grid bias of V is the
potential between the slider of R4 and the earth line plus any
A.V.C. bias, while the H.T. supply for this valve is the voltage
between the slider of R4 and the diode cathode. The total
voltage across R4 is usually about 40-80 volts.
Negative Bias on the Diode Anode
Now it can be seen that if the voltages on the muting valve
are such that it passes no anode current, it has no effect on the
operation of the receiver. If the voltages are such that the
valve does pass current, however, there will he a voltage drop
across- Ri, which is the diode load resistance. The anode of V
will consequently be negative with respect to the diode cathode
and as the diode anode is in connection with it, it will be
negative with respect to its own cathode. The diode anode is
thus biased negatively and rectification cannot occur as long as
the signal is smaller than the bias.
In normal operation, the slider on R4 is adjusted in the
absence of any signal to such a position that the detector is
rendered just inoperative. The valve V is then passing just
enough current to bias the diode by 1-3 volts. When a signal
is tuned in, V receives an increase of grid bias from the A.V.C.
system, and if it be large enough its anode current will cease
and the bias on the detector will disappear so that correct
operation is secured.
If the A.V.C. bias is not great enough completely to cut off
the anode current of V, the detector bias will be only partially
removed, with the result that the detector will operate only
inefficiently and with severe distortion.
It can be seen that for any setting of R4, there is a range of
signal strengths over which the muting is only partially realised
178
AUTOMATIC VOLUME CONTROL SYSTEMS
Fig. 85 1 A simple method of obttining Q.A.V.C. with s diode detector
and over which distortion occurs. This is inevitable and the
aim in design is to make the range as small as possible and to
make it occur only on weak signals which are not usually^ of
much importance. Q.A.V.C. is consequently a fitting which
is of doubtful value in all but very sensitive receivers.
The faults which can arise in an arrangement such as that of
Fig. 85 are very few and are really limited to the D.C. voltage
supplies, which can be readily checked with a voltrneter, the
valve itself, and the A.V.C. feed. In certain modifications,
decoupling is included in the anode circuit of the muting valve
and there is, of course, then the additional possibility of a
defect in these components.
There are many other Q.A.V.C. systems employed and they
are too numerous to be treated in detail here. They all operate
upon the same basic principles, however, and it is usually
quite easy from an inspection of the circuit diagram to see how
they work and to devise suitable testing methods for any
particular case.
179
CHAPTER 17
SHORT-WAVE RECEIVERS
D issatisfaction is often expressed with short-wave
apparatus of the best design and in good condition,
because the user fails to realize the different con-
ditions existing on the short and the ordinary broadcasting
bands. Filled by the expectation of enjoying programmes
from the Antipodes, to say nothing of America, the new-
comer to short-wave reception is often severely disappointed
and believes the equipment to be faulty, but as often as not
there is nothing wrong with the receiver.
It must be realised, first of all, that the chief programmes
broadcast by other countries occur, like ours, in the evening.
Eight o’clock in America, however, is roughly i a.m. in this
country, and the same hour in Australia corresponds to eleven
o’clock in the morning. It usually happens, therefore, that
during the normal listening hours, the programmes from distant
stations are not at their best, and many stations are not trans-
mitting at all. Before trying to receive a particular station,
therefore, it is wise to consult a list of stations to make sure
that it is working.
In the second place, one must not forget that the effectiveness
of different wavelengths varies with the time of day and year.
In general, reception above 30 metres is better after dark than
during the daylight hours, but the converse is generally true
of wavelengths below 20 metres. The band between 20 metres
and 30 metres is often at its best around sunset.
These figures for wavelengths are naturally very approximate
and considerable variations are often experienced. Freak con-
ditions occur when totally different results are obtained, but on
the average they give a good guide to what one may expect.
Variable Reception
This question of wavelength is well worth bearing in mind
when it is desired to hear a particular programme from, say,
America. It often happens that the same programme is
broadcast from several different stations and the one which is
received best depends largely upon the time of day. During
the daytime, the American stations on about 16 metres are
generally the best heard. Towards evening stations around
19 metres begin to come in, and are often very good indeed for
180
SHORT-WAVE RECEIVERS
about an hour before and after sunset. Sometimes good recep-
tion will continue throughout the evening, but at others it is
necessary to go to higher wavelengths for good results once
darkness has fallen.
In addition to this question of wavelength it is necessary to
realise that conditions on short waves change very greatly and
very rapidly. *At one time, it may be possible to receive Ameri-
can stations with a degree of quality and volume suggestive
of the best Continental transmissions on the medium waveband ;
a few hours later conditions may have changed so much that
no Transatlantic stations are audible, or fading may set in which
is so severe that the programme becomes unintelligible.
This may seem rather a black picture of short-wave reception,
but the author wishes to emphasise the disparity between the
conditions of ordinary broadcast reception and those existing
on short waves. There is no doubt whatever that at times
reception of entertainment value can be obtained from the
most distant stations, and most days intelligible reception is
possible from at least a few American stations provided that
the time and wavelength be properly chosen. Reception is,
however, unreliable to a degree almost unknown on the longer
wavelengths. When testing short-wave equipment, therefore,
it is not always easy without an extended test to determine
whether or not the apparatus is functioning correctly, and it
is always advisable when possible to have another receiver
available so that a direct comparison can be made.
Like all other receivers, short-wave sets can be divided into
the two categories of straight sets and superheterodynes. The
latter are now by far the more widely used, particularly with
medium and long wave ranges incorporated, when they are
really all-wave sets. The straight set is by no means dead^
however, and many are still in use. It is almost impossible to
obtain any useful degree of selectivity at signal frequency with-
out making full use of reaction and, in general, this full use of
reaction cannot be realized with a tuned R.F. amplifier. The
majority of S.W. straight sets, therefore, are of the det.-A.F,
type, but some have one R.F. stage. They rely chiefly or^
reaction both for their sensitivity and for their selectivity. It
is, therefore, very important to obtain reaction which is smooth
and free from overlap. This is not always easy, but it is essential
if the receiver is to be of any real use.
In order to obtain satisfactory results, the tuning condenser
must be of high quality and fitted with a good slow-motion
dial free from any trace of backlash. For a purely short-wave
181
WIRELESS SERVICING MANUAL
receiver, the capacity of the condenser should not exceed
0*000 1 6 fJtF. and for an all-wave set, in which some compromise
between conflicting requirements is inevitable, 0*00036 /xF.
The reaction condenser should be equally good mechanically
and of the air-dielectric type with a capacity of some 0*0002 mF,
it is best fitted with a reduction ratio drive.
Now when the components are satisfactory, from a mechanical
point of view, the smoothness of reaction depends entirely upon
the electrical constants, and the valve, the circuit, and the
operating voltages are all important. A typical detector circuit
is shown in Fig. 86 ; it may be followed by any conventional
A.F. amplifier, and the choice of a suitable valve is quite
important.
The type of valve has very little effect upon the efficiency
of detection, for this is almost entirely a matter of the grid
circuit. As far as detection is concerned, the grid and cathode
form the only important electrodes and the action is that of the
conventional diode detector. As a result of detection, A.F.
potentials appear on the grid of the valve, and for these the
valve acts as a conventional A.F. amplifier. Since there is little
Fif. 86 : The detector circuit of a typical short-wave receiver
182
SHORT-WAVE RECEIVERS
difference between the grid circuit characteristics, as regards
theii effect on the efficiency of detection, of valves of widely
different triode characteristics, it would seem that the best
valve to choose would be the one which will give the greatest
gain as an amplifier. In S.W. receivers only small signal
voltages are involved, so that this would be true if reaction had
not to be considered.
The R.F. potentials applied to the grid affect the anode
current, and by means of the reaction coil a suitable amount
of energy is fed back to the grid circuit. The effect is very
nearly the same as if the R.F. resistance of the tuned circuit
were reduced. As reaction is increased the effective resistance
falls and signal strength increases, but stability is maintained as
long as the total resistance is positive. As soon as the resistance
becomes zero or negative, oscillation commences.
When receiving telephony, the resistance should always be
positive, but for weak signals as near to zero as possible. Very
high gain is obtainable by means of a correctly adjusted reaction
circuit, and the importance of correct adjustment will be
realised when it is said that although it is possible to obtain
amplification of several thousand times in this way, a badly
adjusted circuit may give a gain of no more than ten times.
Influence of Reaction
The influence of reaction on the choice of the valve now
becomes apparent. If the valve is chosen with a high ampli-
fication factor to give maximum A.F. amplification, the best
valve is unlikely to give more than three times the gain of one
of lower mutual conductance. It may happen, however, that
the valve which is apparently the poorer enables smoother
reaction to be obtained, and this permits one to approach the
oscillating point more closely and reaction is more effective.
The value of reaction might easily be increased ten times, and
this would more than offset the lower A.F. amplification, and
the apparently poorer valve would give the better results.
In general, the detector valve should have an A.C. resistance
of 10,000-15,000 ohms and a moderate mutual conductance —
i-i *5 mA/V is ample. The anode voltage should not be high,
and in most cases 60 volts is sufficient ; in some a lower voltage
is better but it should always be adjusted by trial and error.
The grid leak and condenser are also important. The con-
denser does not normally require critical adjustment and a value
of 50-100 (X(jiF is satisfactory, but the grid leak should be of
fairly high value. It is not, however, the value of the grid
183
WIRELESS SERVICING MANUAL
leak which is so important as the potential at which the grid
is maintained and this depends not only on the grid les^ but on
the point to which it is returned. In general, and within lirmts,
the higher the resistance of the grid leak and the less positive
the point to which it is returned, the smoother the operation of
reaction.
The best results are usually secured with a grid leak of i~2
megohms returned to the slider of a potentiometer connected
across the filament (Rz, Fig. 86). When the slider is at the
positive end reaction will usually be “ ploppy,'' but it will be
quite smooth at the negative end. For the greatest sensitivity,
the slider should be set as near to the positive end as possible
while keeping reaction quite smooth. This usually means that
the slider will be set at about one-quarter to one-third of ite
travel from the negative end. Where no such potentiometer is
fitted, the grid leak must be taken to the positive side of the
valve filament, and to obtain smooth reaction a higher value
of grid leak will often be needed, usually 2-5 megohrns.
When using indirectly heated valves no positive vbi^ is
needed, for grid current flows until the grid is more negative
than about i volt. Owing to this, it is often necessary with
such valves to apply a small negative grid bias of perhg^Ds 0-5
volt. This is most easily done by adopting the. connections
of Fig. 87, in which R2 is the bias resistance. It should be
adjustable and can have a maximum value of some 500 ohms.
Adjustment is carried out in the same way as with tho poten-
tiometer of a battery set, bearing in mind that an increase in the
value of the resistance corresponds to moving the slider of the
latter towards the negative end. ' ^
It often happens that
threshold howl prevents |
proper reaction effects
from being secured,
for when reaction is
adjusted to the oscilla-
tion point a growl or
howd appears. There
are several causes ol
this trouble, but one of
the commonest and the
most often overlooked
is the nature of the
anode circuit load
impedance at low
SHORT-WAVE RECEIVERS
frequencies. In order to avoid threshold howl this must not
be highly inductive. When transformer coupling is used,
the primary is an inductance and threshold howl is
common. It sometimes happens, of course, that no trouble is
experienced, probably owing to the primary being damped by
the losses in the transformer, but with good quality modern
transformers it is very likely to occur. The remedy is to shunt
the transformer primary with a resistance of some 10,000-
20,000 ohn\s (shown dotted at R3 in Fig. 86), and thus damp
it artificially. This resistance naturally lowers the A.F. ampli-
fication, but just as in the case of the valve, the improvement
in reaction more than outweighs the loss in most cases.
R.F, Currents in the A.F. Circuits
The second common cause of threshold howl is merely the
presence of R.F. currents in the A.F. circuits and this is a
prevalent cause also of hand-capacity effects, particularly when
telephones are used. The remedy is to use adequate filtering
after the detector. The condenser C2 should always be included
and can usually be about o*ooi (xF. When the resistance R3 is
used, a larger capacity can be employed without affecting the
quality of reproduction and C2 can then be about 0*002 [xF,
and sometimes as high as 0*005
In many cases, a resistance in place of the choke Ch will
give better filtering. A suitable value would be 20,000 ohms,
but as the anode current flows through it, a higher H.T. voltage
will be needed, and it may have an effect upon the smoothness
of reaction.
When the circuits have been correctly adjusted for reaction,
it may be found that although results are satisfactory over a
portion of the tuning range there are dead-spots at w^hich the
reaction has to be greatly advanced before oscillation can be
secured or at which no oscillation at all can be obtained. There
are two causes of this — the aerial and the R.F. choke Ch. If
the latter be responsible, the trouble will persist with the aerial
removed, and the remedy is either to use a better choke or to
replace it by a resistance, with which the effect does not occur.
With a circuit such as that of Fig. 86, where the aerial is
coupled to the circuit to which reaction is applied, dead-spots
are inevitable w*hen the aerial coupling is tight. They occur
when the tuned circuit is resonant at harmonics of the natural
frequency of the aerial, and the remedy is to loosen the aerial
coupling. It will then be too loose for the best results at other
G 185
WIRELESS SERVICING MANUAL
frequencies, so that with a receiver of this nature the aerial
coupling should be adjustable by means of a panel control.
This naturally complicates the tuning, and it is consequently
common practice to include an R.F. stage largely for the purpose
of isolating the tuned circuit from the aerial. Dead-spots
cannot then occur and one rather troublesome control is
avoided ; furthermore, the calibration of the set is unaffected
by the aerial, and radiation from the aerial is reduced if the set
is operated in an oscillating condition, while the stage may give
some small degree of amplification.
The Aerial Circuit
The receiver then takes the form of Fig. 88 and is a very
considerable improvement over the simpler set. It might be
thought that it would be advantageous to tune the aerial circuit,
but this is not always the case. In the first place, ganging
would be difficult unless the aerial coupling w'ere loose, for the
same effect that produces dead-spots with the simple detector
circuit would cause mis-ganging at certain wavelengths.
Secondly, although such troubles could be avoided by using a
separate tuning control, this would render operation more
difficult. Thirdly, serious difficulty from instability might be
found if the R.F. stage gave any useful degree of amplification ;
and fourthly, reaction would probably be less effective. This
last is the most serious drawback and is a defect almost inherent
in any receiver embodying both R.F. amplification and reaction.
As reaction is increased in such a set, the R.F. stage comes
nearer to a state of instability, and the R.F. valve may start
oscillating before the detector. The amplification thus becomes
critically dependent upon the operating conditions of both R.F.
and detector valves and upon the precise tuning of all circuits.
Tuning then becomes very difficult if the highest sensitivity
is required.
This does not, of course, mean that a tuned aerial circuit is
impossible, but merely that it is not usually worth while, and
the arrangement of Fig. 88 is consequently generally adopted.
I'here is only one difficulty which is likely to occur with this
arrangement and this only when the set is used near a medium-
wave broadcasting station. Cross-modulation may then occur
and the programme of the local station be superimposed on
every station received, although in the absence of a signal no
trace of the local station will be evident. This trouble is usually
a sign that the aerial choke Ch i is of too high an inductance and
is resonating the aerial circuit in the broadcast band. The
186
SHORT-WAVE RECEIVERS
use of a choke of lower inductance will usually effect a cure
without affecting the performance on short waves.
The second class of short-wave or all-wave receivers — the
superheterodyne — is really no different from its medium- and
long-wave counterpart. It can develop the same troubles and
the same remedies apply. It is, therefore, hardly necessary to
treat such sets here, for the information given in other chapters
dealing with testing applies equally to this case. The design
of short-wave superheterodynes is, of course, quite another
matter, but that is a subject which cannot be treated here.
There are, however, certain features of some short-wave
superheterodynes which deserve mention, for although they are
not inherently peculiar to short-wave sets, in practice they are
rarely found in any other set. Short-wave sets of the so-called
communication type, for instance, often include a beat-frequency
oscillator and sometimes a crystal-filter in the I.F. amplifier.
They are both used primarily for the reception of C.W. Morse
signals.
Beat-Frequency Oscillator
The beat-frequency oscillator consists simply of an oscillator
187
WIRELESS SERVICING MANUAL
tunable by a panel control over a range of some ± 3 kc/s about
the mid-band frequency of the I.F. amplifier. It is coupled
in some way to the I.F. amplifier, but the coupling is not usually
adjustable.
In addition to the panel control, a trimmer is generally
provided for adjusting the frequency. This is adjusted while
aligning the I.F. amplifier. When the other adjustments
have been raade and with the test oscillator giving a fairly
small signal, switch on the B.F.O., set its panel control to the
mid-point of its range, and adjust the trimmer to zero beat.
I'hen swing the panel control on either side of its mid-point and
check that the beat notes at each limit of travel are about the
same.
If they are not the same, set the control a little off zero beat
on the side giving the greater range, and readjust the trimmer
for zero beat. Having found in this way the electrical mid-
point, the setting should be marked on the panel, for it is the
correct setting for the B.F.O. control while tuning the set.
Defects in the oscillator itself are likely to be no different
from those in any other oscillator. Screening is important,
however, for if it is inadequate harmonics may reach the input
circuits and give rise to spurious signals. On short waves,
how^ever, the order of harmonic involved is so high that trouble
is unusual.
The output of the B.F.O. is important, for it is necessary
that the voltage applied by it to the detector be larger than the
biggest signal voltage. If it is less than the signal, the A-h".
output will depend on the B.F.O. voltage and not on the signal.
Now this requirement introduces complications when the
detector is designed for large signals. For instance, if the
normal detector input is 10 volts or more — the A.V.C. delay
being 10 volts — the B.F.O. injection must be at least 15-20
volts. Even in the absence of a signal, this wdll produce 5-10
volts A.V.C. bias, wnth the result that the set will be very
insensitive. In order to overcome this, A.V.C. is switched off
in some sets when the B.F.O. is brought into use. Unless the
oscillator stability in the frequency changer is very good, this is
not as great a drawback as it sounds, for in C.W. reception
even a trace of frequency instability with A.V.C. is likely to be
ver>' noticeable.
When A.V.C. is desired, however, and it is not possible to
resort to the complication of using a separate A.V.C. amplifier
fed from the last I.F. valve grid, so that the B.F.O. can be
injected at the detector without affecting A.V.C., a compromise
188
SHORT-WAVE RECEIVERS
is adopted. The A.V.C. system is given little or no delay, the
detector is designed for a small input — i or 2 volts only — and
the B.F.O. injection is made quite small. It is made of the
same order or less than, the delay voltage. The B.F.O. then
reduces the sensitivity by only a small amount, or not at all,
and the performance on weak signals, which produce a voltage
at the detector less than that of the B.F.O., is good. Strong
signals, however, produce little or no greater output, and some-
times give somewhat less. In fact, the low B.F.O. injection
produces something of a limiting action.
The- value of the injection is easily checked and in the same
way as the amplitude of the oscillator injection in a frequency-
changer. In fact, the B.F.O. and the detector together actually
form a frequency-changer. With no signal, connect a meter in
series with the diode load resistance and note the change of
current when the B.F.O. is switched on. The injection is
about I *2 times the product of the current change and the load
resistance.
With small injection no serious difficulties arise, but in cases
where it is large, it is often difficult to get sufficient coupling
without making it so tight that the I.F. and B.F.O. circuits
pull each other. Some form of electron-coupled B.F.O. is
sometimes adopted to avoid this, but an alternative is to couple
the B.F.O. very loosely indeed to the grid of the last I.F. valve.
One very convenient arrangement for small injection is to
use a duo-diode-triode for detector and B.F.O. The diodes
act as the detector and the triode as the B.F.O., while the valve
capacities provide the coupling.
Crystal Filters
Another, but somewhat less common, feature of a communi-
cation receiver is a crystal filter. The precise circuit details
differ from set to set, but the arrangement is generally on the
lines shown in Fig. 89. The differences lie chiefly in the form
of the crystal output circuit. In Fig. 89 it i^ apparently the
resistance R, but if this is of high order, it is actually the input
capacity of V2, and R is merely a grid leak. Sometimes a tuned
circuit is used and then R is unnecessary. R is also sometimes
replaced by the primary of a transformer of which the secondary
feeds V2.
Whatever the precise details, the method of adjustment is
much the same in all cases. Strictly, the adjustment of the
crystal-filter is part of the normal operation of the set and not
peculiar to maintenance, and the only control C2 is, in fact, a
189
WIRELESS SERVICING MANUAL
panel control. The presence of the crystal does, however,
complicate the I.F. alignment and it is necessary to deal with
this in some detail.
Before doing so, it may be as well to describe how the crystal
is used in operation, for it is not such a common circuit that it
will be faniiliar to all. First, a word of warning. If the crystal-
filter appears to do little or nothing on its first trial, do not
immediately conclude that it is defective. Everything depends
on the adjustment of C2, and some little skill is needed to
adjust this control properly.
The tuned circuit LCi, the crystal, and C2 are in the form
of a bridge in which Cz is the variable element for balancing.
It is normally adjusted so that it balances the shunt capacity of
the crystal holder. Apart from the piezo-electric properties of
the crystal itself, the bridge is balanced and there is no output
at any frequency. The crystal, however, behaves as a series
resonant circuit of very high Q. At resonance it behaves as a
resistance of moderate value, the bridge is unbalanced, and at
this frequency the signal is passed to the output. For quite a
small frequency difference from resonance the crystal becomes
substantially a high reactance, the bridge is nearly balanced,
and the output very small. The effective band-width is very
small and depends upon the crystal itself as well as upon the
output circuit. It is usually of the order of 50 c/s to 500 c/s.
It will be clear, therefore, that the circuit is not very suited
to telephony, and it is intended primarily for C.W. Morse
reception. It can be used for telephony, however, particularly
if the balance of the bridge is upset slightly in the manner to
be described later.
If the condenser Cz is not set to the right value for balancing
the bridge, the crystal will have very little effect. The best
thing to do then is to set Cz for minimum background noise,
since this usually occurs with minimum band-width and this
in turn occurs when the bridge is balanced.
Even if the bridge is still not quite balanced, it should now
be possible to determine the setting of the tuning control
corresponding to the crystal frequency, for when tuning
through a signal a small “ bump ** should be heard as the inter-
mediate frequency passes over the crystal frequency. By
taking this as a guide, it is easily possible by small adjustments
to Cz so to balance the circuit that the signal disappears except
at the one critical setting of the tuning control.
Signals should now tune-in very sharply, and telephony will
be very deep toned ; if the filter is of a very selective type,
190
SHORT-WAVE RECEIVERS
speech may well be unintelligible. There will also be a con-
siderable apparent loss of sensitivity. It is a real loss on tele-
phony, because of the severe sideband cutting, but it is only
an apparent loss on C.W. and occurs through the drop of
background noise.
One peculiarity of the high selectivity should be noted. It
is imperative to tune-in a signal with the B.F.O. at the same
frequency as the crystal resonance. The signal should be
tuned to zero beat, and then the B.F.O. adjusted to give the
required beat note. If one tries to tune wdth the B.F.O.
already ofl'set, tuning will be confusing. It may be impossible
to tune to zero beat, there will be one setting only of the tuning
control at which the signal is audible and this may not coincide
with the correct setting for maximum signal with the B.F.O.
properly adjusted.
If there is any difficulty in setting the B.F.O. to the crystal
frequency, tune-in some signal and adjust the B.F.O. control
wffiile rocking the tuning control slightly until the combination
of settings giving the loudest beat note is found. The signal
is then properly tuned to the crystal, and by turning the B.F.O.
to zero beat it can also be set to this frequency. It is a good
plan to mark the setting.
Unbalanced Filter
The balancing control, or phasing control as it is sometimes
called, is a panel control because it is often useful to work with
the bridge slightly unbalanced, d^hc resonance cur\e is then
asymetrical, but has a very deep crevasse on one side. By
tuning so that this crevasse falls on an interfering signal, it may
often be eliminated. If interference is found, therefore, it is as
well to try unbalancing the bridge slightly, and it will usually
be possible to find a critical setting of Cz at which the inter-
ference disappears. This is especially useful in telephony
reception, for even a bad heterodyne whistle can be eliminated.
On account of the unbalanced condition of the crystal, the
frequency response is usually good enough for intelligible
telephony. There is often some amplitude distortion, however,
because the asymetric resonance curve makes reception almost
single-sideband.
Turning now to I.F. alignment, the important thing is to
make sure that the I.F. circuits are so adjusted that the mid-
band frequency is the same as the crystal resonance frequency.
In all other respects it is entirely normal.
191
WIRELESS SERVICING MANUAL
Fig. 89 ; For extreme selectivity a quartz crystal is sometimes used. The
capacity of its holder is neutralised by adjusting C2
The nominal crystal frequency is usually known, and its
actual frequency is unlikely to differ from it by more than 2
kc/s. It will, therefore, be within the pass-band of an l.F.
amplifier adjusted to the nominal frequency. If the accuracy of
calibration of the test oscillator can be relied upon to within
2 kc/s at the intermediate frequency, the best course is to align
the l.F. amplifier roughly to it. I'hen bring the crystal into
circuit, adjust the balancing condenser, and tune the test
oscillator to the crystal. It will usually be necessary to use the
test oscillator without modulation and to have the B.F.O. on
when doing this. Nearly all test oscillators, and many expensive
signal generators, have too much incidental frequency modula-
tion for them to be of any use in the modulated condition with
sharp crystal filters.
Having got the oscillator set to the crystal frequency, normal
l.F. alignment procedure applies. Circuits associated with the
crystal filter are usually adjusted simply for maximum output
with the crystal in circuit and in the balanced condition.
There are other types of crystal filter to that shown in Fig. 89,
but they are not often met. They are chiefly more complex
structures designed to give a band-pass effect. Some of them
are quite difficult to adjust and need special equipment. In
view of their rarity, they need not be discussed here.
192
CHAPTER 18
SHORT-WAVE CONVERTERS
A lthough not commonly used now, short-wave con-
verters were quite popular a few years ago and they are
still met at times. A properly designed converter used
with a good receiver is capable of giving a very fine performance
indeed, but poor apparatus is naturally unsatisfactory.
Unfortumtely, a good converter is not cheap and many employ
types which suffer from inherent defects which it is rarely
possible to cure without radical alteration. When dealing with
such apparatus, therefore, one must be careful to distinguish
between those faults which are inherent to the design and those
which are the result of a defective con)ponent. The latter are,
of course, no different from those which mav occur in other
equipment and can be found by the application of the usual
testing methods.
One of the commonest arrangements used in the older con-
verters is a single valve, often a triode but sometimes a tetrode
or pentode, connected to function as an oscillator. The only
tuning control is the variable condenser controlling the oscillator
frequency, so that the system has the advantage that no ganging
is required.
From the performance point of view, however, the arrange-
ment is not good, for there is no protection whatever against
second -channel interference nor from any of the other forms
of interference peculiar to the superheterodyne. Every station
can be received at two different dial settings, in consequence.
The efficiency is also generally of a low^ order and the signal to
noise ratio not very high.
The greatest fault of all, however, and it is the greatest
because it annoys not the owner but his neighbour, is the very
serious degree of radiation from the aerial which occurs with
converters of this type. The amount of interference w^hich can
occur on the medium waveband from an oscillating detector set
is well known, but it is not generally realised that the autodyne
converter is in no way better. Actually, it is often much
w^orse, and serious interference with other short-wave receivers
within a radius of ten miles or so may occur.
More modern types of converter often use a heptode, octode,
or triode-hexode valve for frequency-changing. The triode
section functions as the oscillator, but is separated from the
aerial circuit by the internal screening of the valve. Provided
193
If short-wave reception is to be anything nriore than a play-
thing, it is essential for signal-frequency tuned circuits to be
included, for only in this way can the interference troubles be
avoided. The type of converter employing the triode-hexode,
heptode, or octode and one signal-frequency tuned circuit is
capable of a much better performance. The efficiency and the
signal-noise ratio are both considerably increased ; there is
some discrimination against second-channel interference, but
not a great deal. The selectivity of the signal circuit is great
194
SHORT-WAVE CONVERTERS
enough to remove most other sources of interference, however,
so that in practice there is a big improvement.
Unfortunately, the addition of such a circuit considerably
increases the cost of the converter, for not only have additional
coils and a gang'condenser to be provided, but the complications
of ganging ensue. Nevertheless, for serious short-wave listen-
ing, signal-frequency tuning is essential and for the best results
two tuned circuits are needed in addition to the oscillator. An
R.F. valve is then included and there is a further very con-
siderable gain in amplification and in signal-noise ratio. With
two good tuned circuits, second-channel interference can be
kept at a very low level provided that the intermediate frequency
is not lower than 400 kc/s.
Parasitic Oscillation
I'he commonest troubles encountered in converters of this
type are parasitic oscillation and pulling. Parasitic oscillation
often occurs in the oscillator circuit and then the remedies
described in pages 82-84 will usually ehect a cure. It can
occur in the hexode or heptode section of the valve, however,
atjd the remedy is then to insert a resistance of about 50-100
ohms in the control grid lead. The remarks on pages 157-158
about the amplitude of the injected oscillator voltage should be
borne in mind, however, for this is just as important in a
converter as in a complete short-wave receiver.
Pulling is usually most evident when the signal and oscillator
circuits arc separately tuned and, as its name implies, the
setting of the oscillator control becomes dependent on the
setting of the other. With such interaction between the
controls, tuning is difficult. It is not always possible completely
to eliminate it ; in fact, it is theoretically possible to avoid it
only when the coupling to the oscillator is purely electronic and
this is never completely achieved in practice.
In the author’s experience there should be no noticeable
pulling with the triode-hexode frequency-changer on wave-
lengths above some 10 metres and only slight traces down to
5 metres. With heptodes and octodes rather more pulling is
likely to be experienced below 20 metres, and it is often serious
on the ultra-short waveband.
Any pulling more serious than that indicated must be taken
to mean that unwanted couplings exist betw^een the signal and
oscillator circuits external to the valve. Every endeavour
should be taken to eliminate them, therefore, by careful attention
to detail in screening, wiring and decoupling.
When all tuning controls are ganged the effects of pulling are
195 .
WIRELESS SERVICING MANUAL
not so evident, but they make the operations of ganging difficult
to perform. Sometimes, however, pulling makes tuning
troublesome by causing a curious effect. As the control is
rotated across a station the strength of the signal increases in
the usual way, but instead of reaching a maximum and gradually
decreasing aften^^ards, a point is found beyond which the
slightest further movement causes the strength to drop
enormously. If then the control be turned back, the signal
does not attain the same volume at the previous maximum but
at a setting appreciably different.
The interaction between the tuned circuits is responsible
because under certain conditions a very small change in the
tuning of a signal-frequency circuit suddenly causes a much
larger change in the tuning of the oscillator circuit. The effect
is particularly serious where A.V.C. is used, because the A.V.C.
voltage usually has a slight effect on the oscillator frequency.
This should be too small to be noticeable, even at the high
frequencies used in short-wave reception, but it will not be in
the circurr. stances indicated.
Both pulling and oscillator stability become very important
on short waves and if there is any serious degree of interaction
betsveen the tuned circuits or if the oscillator frequency wanders
with changes in the voltages applied to the valves, A.V.C. is
worse than useless. In the case of the oscillator some drift in
frequency is to be expected for half-an-hour or so after the set
is switched on ; in fact, until the valves have attained their
normal operating temperature. Other factors affecting
frequency-stability are dealt with on page i6o.
Microphonic Howling
The most commonly encountered difficulty in short-wave
reception, and particularly in ultra-short-wave reception is,
howling due to microphony. This is dealt with on page
13 1 for the case of receivers covering only the medium and
long wavebands, but it is much more common and more severe
in sets or converters operating on short waves. It is often
more troublesome in converters than in complete sets in which
the relative positions of the loudspeaker and the S.W. equip-
ment are fixed by the design. Although valves, coils, and even
wiring can be responsible, the trouble is usually due to vibration
of the plates of the variable condenser. Experience shows that
it is imperative to use a flexible mounting for the condenser ;
the degree of freedom necessary in any case, however, depends
largely upon the rigidity of the vanes.
. 196
SHORT-WAVE CONVERTERS
It is not always essential that the condenser be mounted
flexibly on the chassis, for when it is not the chassis itself can
be mounted on blocks of sponge rubber. Owing to the greater
weight of the chassis, however, it is difficult to obtain sufficient
flexibility in this way to meet the requirements of short-wave
reception. Probably the best course is to mount the condenser
flexibly on the chassis and then to stand the chassis on sponge
rubber, thus combining the advantages of both methods.
It should be noted that individual isolation of the condenser is
not always needed. Sometimes the condenser with tuning coils,
waveband switching, and the early valves, are mounted on a
sub-chassis which is joined to the main chassis by an absorbent
mounting. As this sub-chassis is quite light, this scheme is
usually successful.
Ultra-Short Waves
On ultra-short waves, below about lo metres and extending
into the television band, microphony becomes very serious, and
extreme precautions are necessary for its avoidance. This is
because at the higher frequencies involved a given movement
of the condenser vanes causes a much larger change of frequency.
The way in which microphony increases with frequency is
not difficult to understand. Suppose that the I.F. amplifier
has a resonance curve like that of Fig. 90, this is an idealised
curve with a flat top 10 kc/s in breadth. Now if the inter-
mediate frequency is 465 kc/s and the set is tuned to receive a
station on 1,000 kc/s, the oscillator is operating at 1,465 kc/s.
Suppose, further, that acoustic feed-back causes the plates of
the oscillator condenser to
vibrate by an amount such
that the oscillator fre-
quency changes by i
1,000 c/s. The oscillator
frequency then varies
periodically between 1 ,464
kc/s and 1,466 kc/s, and
the intermediate frequency
varies between 464 kc/s
and 466 kc/s. The inter-
mediate frequency thus
varies by i 1 kc — the
same amount as the oscilla-
tor — about the mean value.
Fig. 90 : An idealised resonance curve of a
superheterodyne I.F. ampfifier
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WIRELESS SERVICING MANUAL
Owing to the flat-topped resonance curve of Fig. go, however,
this causes little harm, for ignoring small effects due to the
sidebands if the carrier is modulated, there is no variation in
the detector input. Suppose, however, that we now tune in a
station on lo Mc/s (30 metres). The oscillator frequency will
now be 10,465 kc/s, but the variation due to microphony will
not be dt: i kc/s, for the amount of vibration of the condenser
plates will be the same as before. Assuming the same L/C ratio
in the tuned circuit the percentage change of frequency will be
the same ; that is, the frequency will vary by i part in 1,465,
or 10,465/1,465 = 7*15 kc/s. The intermediate frequency
will consequently sweep between 457*85 kc/s and 472*15 kc/s,
and it will no longer remain within the limits of the flat top of
the resonance curve, but will sweep over the sloping sides,
causing the detector input to vary. This naturally varies the
sound output from the speaker and hence the vibration of the
condenser vanes.
Higher Frequency Drawbacks
At higher frequencies, matters are still worse. Thus, at
41*5 Mc/s (7*23 metres) the frequency used for the sound
accompaniment to television, the oscillator frequency would be
41*965 Mc/s, and on the same basis as above the change in
intermediate frequency would be no less than ± 28*74 kc/s.
In practice, of course, it is very doubtful whether the changes
are actually as great as this, for there are various modifying
factors which have been ignored for simplicity. Nevertheless,
the broad fact emerges that the less selective the I.F. amplifier,
the greater the freedom from feed-back effects. A certain
degree of selectivity is needed for the avoidance of interference
and one cannot avoid microphony with safety by adopting this
means on the ordinary short wavebands.
It is, however, legitimate to do so on the ultra-short wave-
lengths, for here interference is not yet a problem, and the use
of a wide band-width for the I.F. amplifier not only renders
microphony of little importance but it enables really high-
quality reproduction to be obtained, since the highest musical
frequencies can be fully reproduced. For ultra-short-wave
working, a special superheterodyne is thus needed for the best
results, a band-width of 20-30 kc/s with an intermediate
frequency of 5 Mc/s is satisfactory. A converter can be used
with an ordinary broadcast set, but the average set is too
selective to prevent some difficulty being experienced from
microphony.
198
CHAPTER 19
THE LOUD SPEAKER
T he defects which can develop in a moving-coil loud
speaker are quite numerous, and are often very difficult
to trace. Fortunately, however, they occur comparatively
rarely, for the modern speaker is a fairly robust instrument and
will withstand a lot of misuse. Defects in the field winding,
such as an open-circuit or short-circuit, are easily located by a
resistance test and need not be further considered. The faults
which are most difficult to trace occur in the cone and its
suspension system, and they produce distortion which it is
easy to mistake for that introduced by an overloaded valve.
One of the commonest faults is a dirty air-gap between the
poles of the magnet. It must be remembered that the magnet
is powerful, and in the course of time it collects stray iron
filings and these accumulate in the gap, together with dust and
dirt, and foul the moving-coil. Its movement is consequently
impeded in an irregular manner and the quality of reproduction
is affected, becoming “ fuzzy ” on loud passages. This trouble
is rather more likely to occur with a permanent magnet speaker
than with an energized model, since its magnetism is always
present, whereas the full magnetism of the latter is only produced
when the field is energized.
In order to clean the gap properly it is necessary to remove
the cone and moving-coil assembly and to wipe the gap
thoroughly with a piece of cloth wrapped round a thin sliver of
wood. Thorough wiping is essential if all particles are to be
removed, for the particles of magnetic materials have natjarally
a strong tendency to stick to the faces of the magnet, and it is
a great help to smear the cloth with a little vaseline.
On replacing the cone assembly, it is necessary carefully to
centre the coil in the gap so that it moves freely without coming
into contact with the pole pieces. Centring is often needed
even when the cone has not been removed, too, for the continual
vibration to which a speaker is subject may in time loosen the
centring screw or, more often, strain the spider.
When the air-gap is fairly wide, centring is quite easy and is
best done by trial and error. A typical arrangement is shown
in Fig. 91 ; the centring screw should be slackened and the
coil allowed to take up its own position. Then tighten the
screw with the fingers so that it only just holds the spider.
If the cone is now moved gently in and out by grasping it at
199
WIRELESS SERVICING MANUAL
diametrically opposite points with the fingers, it is easy to tell
whether it touches the magnet or not. If it does, determine
which side it is touching and move the spider appropriately.
When the central position has been found, the centring screw
must be properly tightened.
It is very important that all screws and nuts on a loud speaker
be thoroughly tight, otherwise they will rattle, usually on
certain musical notes only. A similar efTect, which varies
from a rattle to a buzz, can be caused by an imperfect seam in
a cone, or a poor joint of cone to spider, cone to moving coil,
or cone to surround. In many cases such joints are made with
overlapping paper cemented together ; if the cement cracks the
two edges will rattle against one another and cause distortion at
certain frequencies and particularly at large volume. The
remedy, of course, is to cement the surfaces together again,
and this is readily done with a cement consisting of celluloid
dissolved in amyl acetate.
Faults of this nature are not a common occurrence with the
ordinary loud speaker and under normal conditions of use ;
they are most likely to occur when the amplifier has a large
output stage. They are, however, very probable if the speaker
be used for any length of time with an amplifier which is motor-
boating, for the large amplitudes developed are a severe strain
upon the cone suspension and the centring device.
There are, of course, many other defects which may be
present in a loud speaker, such as a poor frequency response, or
a non-linear motion of the cone. These, however, are by no
means easy to remedy and are, in fact, often defects in the
original design and not faults which have developed with use.
There is consequently little which can be done to remedy them
and the only certain method of deciding whether any fault is
peculiar to a particular specimen, or whether it is common to
all models of that type, is a comparative test of two specimens
under the same conditions.
Even with the best speaker, the performance will not be good
unless it is correctly used and the baffle or cabinet is a very
important item, as is also the matching of the speaker to the
output valve. From most points of view a flat baffle is the
best and it should be constructed from rigid material if it is
not itself to introduce resonances. Good results can be secured
with a baffle 3 ft. square of 7-ply wood provided that it is braced
and rigidly mounted. Theoretically, of course, a much larger
baffle is necessary for the full reproduction of the lowest
frequencies, but this is rarely practicable. It is sometimes
200
THE LOUD SPEAKER
Fig, 91 : The moving-coil can readily be centred by slacking off the centring
screw and moving the spider so that the coil can move freely
possible, however, to build a loud speaker into a wall between
two rooms so that each room is served by one side only of the
cone, and this constitutes an ideal baffle.
Cabinet Resonance
In most practical cases, however, the loud speaker must be
used in a cabinet and great care must be taken to avoid
resonances. These occur at low frequencies, and while they
give a false bass which is not always objectionable on music,
they often make the reproduction of speech unnatural and
intolerably “ boomy The golden rules for the avoidance of
box resonance are to use as wide and shallow a cabinet as
possible and to construct it rigidly of heavy wood. In practice,
however, it more often happens that one is confronted with a
cabinet of light wood, sometimes ply- wood, having a depth
nearly equal to the width and the problem is so to arrange
matters that box resonance does not occur.
201
WIRELESS SERVICING MANUAL
This is naturally much more difficult, but it is a problem
which often arises. In many cases no complete solution is
possible, but some improvement can usually be made. If
space permits, the best course is probably to line the cabinet
with slag wool, but this is rarely possible when the receiver is
contained in the same cabinet as the speaker, and the best
course in these circumstances is usually to line the cabinet
with some acoustically dead material, such as Celotex board.
If this cannot be done, or proves ineffective, it is advisable to
relieve matters by cutting holes in the cabinet at appropriate
points. As much of the bottom as possible should be removed,
assuming that the cabinet has legs, so that the bottom is spaced
from the floor, and if this proves insufficient holes can be cut
in the sides and covered with gauze to prevent the entry of dust.
Such holes will naturally reduce the baffle area and so tend
to reduce the bass response, but they are very effective in
eliminating box-resonance and the net result is usually a con-
siderable improvement in the quality of reproduction.
Even the best speaker and baffle will not give good results,
however, if the matching to the output stage of the receiver is
incorrect. Most valve makers publish a figure for the optimum
load impedance required by an output valve, and figures for
current types also appear in the Wireless World Valve Data
Supplement, The speech-coil impedance of the loud speaker
can be obtained from its makers and the transformer ratio is
then easily calculated from the formula, ratio = \/Zl/Zs,
where Zl is the load impedance required by the output stage
and Zs is the impedance of the speech coil. When Zl is greater
than Zs, as it is in practically all cases, the ratio is a step-down
from primary to secondary.
Load Impedance
In general, when two output valves are used in parallel in
the output stage the load impedance required is one-half of
that needed by one valve, while when they are in push-pull the
load impedance should be twice the figure for one valve. It
should be understood that the valve maker’s figures for load
impedance apply only when the valves are used under the rated
conditions. Set designers often choose different operating
conditions, particularly when a push-pull output stage is used,
and the load impedance then required may differ from the
valve makers’ figure.
With triode output valves, the matching is not very critical
and appreciable departures from the optimum conditions can
202
THE LOUD SPEAKER
be made without seriously affecting the performance. The use
of a transformer of higher ratio than the optimum leads to some
drop in sensitivity but may actually improve quality slightly.
A lower ratio than the optimum, however, may give increased
sensitivity but is likely to affect the quality seriously. With
pentode output valves, however. Class “ AB ” and Class “ B
types, very little mismatching can be tolerated from the point
of view of quality, and serious distortion may occur if the load
on the output stage is not kept quite close to the optimum figure.
The output transformer is important apart from its ratio,
and if it is not to introduce distortion it must have a high
primary inductance and a very low leakage inductance. These
requirements are mutually opposing and a good transformer
will consequently have a large core, many turns, and the
primary and secondary windings will be in many interleaved
sections. Except with small output stages, a large core is
necessary to avoid amplitude distortion due to saturation of the
core at low frequencies.
The important factor is the ratio of the primary reactance at
the lowest frequency to the effective circuit resistance, which
is the output resistance of the amplifier in parallel with the
load resistance required by the amplifier.
For the best results with an output of 4 watts, the core
section should not be less than i *5 sq. in. using Stalloy lamina-
tions. This applies when the load impedance is 10,000 ohms,
the amplifier output resistance is 1,500 ohms and the primary
inductance 70 H. The use of a lower load impedance would
permit a smaller core to be used or larger power to be handled
by the same transformer.
When a transformer is to be selected for a large amplifier,
therefore, care must be taken to obtain one with adequate
inductance and core if the bass is to be properly reproduced
without distortion, and of low leakage inductance if the higher
frequencies are to be fully reproduced. On account of the
high A.C. resistance of pentodes, the output transformer
would have to be impossibly large if it were not for the fact
that negative feed-back can be used to reduce the output
resistance to any desired figure.
The High- Note Tweeter
With high quality equipment the moving-coil speaker is
sometimes supplemented by a high-note tweeter. In general,
the moving-coil speaker fails to maintain its output at the
highest audible frequencies and an additional speaker, specially
203
WIRELESS SERVICING MANUAL
PIEZO ELECTRIC
TWEETER
92 : The method of connecting a piezo-electric tweeter and moving-coil
speaker is shown here
designed to deal with these frequencies only, enables a con-
siderable improvement in reproduction to be obtained. The
use of a tweeter is, of course, only justified for high quality
local reception, for the avoidance of interference necessitates
the suppression of the highest frequencies when receiving
other stations.
The piezo-electric principle is commonly employed for
tweeters and the best method of connection is usually the one
shown in Fig. 92. The potentiometer is included in order
that the correct balance between the two speakers may be
secured, for moving-coil types differ widely in their sensitivity.
It is important to note that if the correct effect is to be secured
the tweeter must be mounted close to the moving-coil speaker,
and it is usually best just above it.
204
CHAPTER 20
EXTENSION LOUD SPEAKERS
T O everyone there comes at some time the problem of
fitting one or more additional loud speakers to a receiver.
Many modern receivers are, of course, arranged by the
makers so that this is easy, and there is then nothing to do but
follow their instructions. In the general case, however, it is
necessary to exercise some care if satisfactory results are to be
secured .
The correct solution to the problems involved may turn out
to cost more than one at first expects, so that it is best to tackle
the question from two different angles. One way is to consider
the matter from the scientific point of view and find the arrange-
ment which, regardless of cost, will give the best results. This
will often be one the cost of which is only justifiable in a high
quality permanent installation. The second method is to
adopt a more commercial attitude and find the cheapest system
which will enable reasonably good results to be obtained.
Considering this second method first, the usual procedure is
to obtain an additional permanent magnet type speaker complete
with output transformer and to connect the primary of this
transformer in parallel with that of the existing output trans-
former. Since the extension leads will then usually be joined
to positive H.T., it is a good plan to modify the system by
inserting a 2 fiF condenser in series with each lead at the
receiver end. An accidental short-circuit to earth from any
point on the extension line will not then cause damage through
short-circuiting the H.T. supply.
It is easy to see that the drawback of this simple scheme is
that the operating conditions of the output valve are upset, for
its load impedance is lowered. If the valve normally works
into the correct impedance its load will be too low when both
speakers are in use, and amplitude distortion will be increased.
Furthermore, the extra speaker robs the other of power, so that the
volume control must be turned up to maintain normal volume,
and there is again the likelihood of an increase in distortion.
It often happens, however, that a much lower level of volume
is required from the extra speaker, since in many cases it is
used in a smaller room than the main one. It is then possible
to improve matters considerably by deliberately mismatching
the extra speaker. If the output valve requires a load impedance
of 5,000 ohms, for instance, which is provided by the main
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WIRELESS SERVICING MANUAL
speaker, then instead of using an extension speaker having a
transformer to give this same value of load, provide it with one
giving an impedance of, say, 20,000 ohms.
Advantage of Mismatching
Instead of the load on the valve decreasing from 5,000 ohms
to 2,500 ohms, it will then only fall to 5,000 X 20,000/25,000 =
4,000 ohms, and the operating conditions will be much less
affected. Furthermore, instead of the power being divided
equally between the speakers, it will divide in inverse proportion
to the transformer primary impedances. Thus, in the above
case, the main speaker will take four-fifths of the power and the
extra speaker one-fifth ; the difference in volume between the
two will consequently be 6 db. Ignoring any loss due to the
change in load impedance on the output stage, the volume of
the main speaker will drop by i db. only, instead of 3 db., when
the extra loud speaker is connected. The volume from the
extra speaker will be 3 db. below that given by equality of
transformer primary impedances.
When this course is adopted it is possible to include a make-
and-break switch in series with one of the extension leads to
permit the extra speaker to be switched on and off at will, and
the performance of the main speaker is hardly affected. Where
a number of extra speakers is required, however, this arrange-
ment is hardly practicable and it becomes necessary to adopt a
more scientifically-designed arrangement.
To take the most general case, let there be a total of ?t speakers
and let it be required to feed each with the same amount of
power. If all speakers are identical their speech coils can all
be in parallel, the extension leads then being in the output
transformer s econdar y circuit. The ratio of this transformer
is equal to \/nZLlZs when Zl is the optimum load required by
the valve and Zs is the impedance of the speech coil of one
speaker.
When the impedance connected to any one line is of the
order of 50-500 ohms, this arrangement is satisfactory for lines
of up to some 200 ft. By the impedance connected to the line
is meant Zs/wi where is the number of speakers connected
to that line. Thus with Zs = 100 ohms, it is possible to feed
two speakers through one line quite safely, provided that
heavy gauge conductors are used, but if there are more than
two speakers extra lines should be used. When the line length
is shorter, however, a lower impedance termination is possible and
it is possible to use 30 ft. or so with Zs no more than 15 ohms.
206
EXTENSION LOUD SPEAKERS
Secondary Extensions Inadvisable
Many speakers have lower impedances than this, however,
and 2*5 ohms is a common value. It is then inadvisable to run
extension leads in the secondary circuits and each speaker
should be provided with its own transformer.
This must be done when the speakers have differe nt values
of Zs and the ratio of each transformer should be \/nZhlZSy
Zs being the speech coil impedance of the speaker under con-
sideration. It will be seen that this gives the same ratio as
before when w, Zl and Zs are the same. Generally, however,
the transformers will not be the same.
With only a single transformer, the primary reactance should
be twice Zl at the lowest important frequency. When each
speaker has its own transformer, however, the primary reactance
of each should be equal to 2nZL ; that is, the primary inductance
in the second case should be n times that in the first. It is not
always possible to adhere to this rule.
The difficulty in fitting each speaker with its own transformer
is that the lines cannot be very long before capacity effects
upset the operation by removing most of the upper musical
frequencies. The speech-coil impedance of the average
speaker is too low to permit a long line in the transformer
secondary circuit on account of the resistance and inductance
of the wires, while the load impedance of the average output
stage is too high to permit a long line being used in the primary
of the output transformer on account of the capacity.
When Long Lines are Used
In cases where very long lines must be used, therefore, it
becomes necessary to take the impedance of the line into account.
Transmission line theory becomes quite complex in all but its
simplest branches, and it will not be treated here. As applied
to extension speakers, however, it involves the use of a trans-
former at each end of the line, one at the receiver to match the
output stage to the line and another at the speaker to match
the line to the speaker.
Where a number of speakers is used and matched correctly,
the output of the amplifier is divided equally between the
speakers. If the output is only sufficient for one speaker,
therefore, it is necessary to redesign the output stage so that it
will supply n times the power.
Not only is an extra loud speaker a common addition to a
receiver, but either a microphone or a gramophone pick-up is
often required. Most sets are arranged for the connection of a
207
WIRELESS SERVICING MANUAL
pick-up and it is necessary only to choose a model of adequate
sensitivity for the particular receiver with which it is to be used
and to connect it to the terminals fitted. Provided that the
connecting leads are screened, trouble is rarely experienced, for
the leads do not usually exceed about three feet in length.
It occasionally happens, however, that it is desired to operate
the pick-up at a considerable distance from the receiver or
amplifier and serious difficulties are then likely to be found.
Pick-ups are normally designed for working into high imped-
ances, o* 1-0*5 megohm, with the result that the capacity between
any considerable length of connecting lead seriously affects the
performance. The capacity is, in fact, very considerable, since
screened leads are essential for the avoidance of hum pick-up.
With the conventional pick-up, there is only one remedy and
this is to connect a step-down transformer between the pick-up
and the leads connecting it to the receiver. With a suitable
transformer, the effect on quality is negligible, but there is a
loss in efficiency unless a step-up transformer is also used to
connect the line to the receiver.
A better course is to employ a low-impedance pick-up which
can be connected directly to the line ; only the step-up trans-
former from line to amplifier is then needed. Pick-ups wound
for an impedance of 600 ohms, however, are not generally
obtainable, probably because a step-up transformer is essential
in all circumstances if their efficiency is to compare with that
of the more common types.
Microphones, too, bring their own problems and with nearly
all types a transformer is needed having a step-up ratio between
the microphone and receiver. The ratio is readily calculated
and is equal to \/Zs/Zp where Zs is the value of the resistance,
usually a volume control, shunting the transformer secondary,
and Zp is the impedance of the microphone. With carbon
type microphones the impedance is practically the same as the
D.C. resistance at the normal operating voltage.
Unsatisfactory results are often secured when a microphone
is connected to the pick-up terminals of an ordinary receiver,
because the amplification is not always sufficient. The output
of a carbon microphone is only about o*i volt and most modern
receivers require an input of the order of i volt at the pick-up
terminals. Other types of microphone give smaller outputs
and in general the better the microphone from the quality view-
point the smaller its output. It is, therefore, often necessary
to connect an amplifier between the microphone and the pick-up
terminals of the receiver if adequate volume is to be secured.
208
CHAPTER 21
THE AERIAL-EARTH SYSTEM
S INCE amplification was difficult to obtain in the early
days of wireless, great attention was then paid to the
aerial and earth system, for improving its efficiency was
the most practicable method of increasing the range of reception
and signal strength generally. With the common use of
modem highly sensitive receivers, however, it is now all too
often neglected ; the listener then pays the price not necessarily
of an absence of signals, but of a poor signal-noise ratio.
Even with a complete absence of local interference there is a
limit to the amplification which can usefully be employed.
As explained on page 8i the receiver itself inevitably introduces
a certain amount of noise and it is essential for the signal
delivered to the set by the aerial to he stronger than this noise
if satisfactory reception is to be secured. The noise level is
fixed for any given receiver and there is very little which can
be done to reduce it ; the level, however, depends on the type
of set and, comparing types of equal sensitivity, will normally
be greater in a superheterodyne in which the first valve is a
frequency-changer than in one which includes an R.F. amplifier.
A set of the latter kind is, in fact, the equal of the straight set
as regards noise.
The actual noise level is not of much importance, however,
if we can obtain an adequate signal-noise ratio. Obviously, it
does not matter whether there is i of noise voltage on the
grid of the first valve or loo provided that when the receiver
is delivering normal volume from a signal the noise is inaudible.
It is always the signal-noise ratio which is important ; the
signal level depends on the power and distance of the trans-
mitter and upon the prevailing conditions. It also depends on
the aerial system, whereas the noise depends only on the
receiver. The signal-noise ratio thus depends on both receiver
and aerial.
Suppose, for example, that two receivers A and B have the
same sensitivity but noise levels of i and lo /xV respectively,
then equal signal-noise ratios will be secured with signal inputs
of, say, lo fiV and loo /xV, and the noise, although audible,
will not be serious. Now if the poorer receiver B is used with
a good outdoor aerial noise will, in general, be negligible on the
stronger Continental transmissions, for at night many of these
provide signals of up to i,ooo ftV. If receiver A is used with
209
WIRELESS SERVICING MANUAL
a poor aerial, such as a small indoor aerial, however, the same
signal may provide no more than one-hundredth the input.
Under these particular conditions there will be a signal-noise
ratio of lo : i for receiver A and i,ooo : i for receiver B ; in
other words, the poorer receiver gives ten times the signal-
noise ratio.
In practice, of course, the difference is rarely as great as this,
but the example emphasises the importance of the aerial.
A good receiver will always give a better performance than a
bad one when the comparison is made on similar aerials, but
it is quite possible for the bad set with a good aerial to be better
than a good set with a bad aerial. After all, a receiver can only
work on the input with which it is supplied and it is the business
of the aerial to supply this input.
Circumstances Governing the Aerial
Now the best type of aerial depends upon circumstances and
there are many factors to be taken into account. In cases
where local interference is present it will probably be necessary
to adopt one of the special lead-in systems described in
Chapter 9, and certain of these are now obtainable in a form
suitable for all-wave reception. In general, anti-interference
aerials are less efficient than ordinary types, so that their use
leads to a lower signal-set noise ratio. The local interference,
however, will be so much greater than the set noise in all cases
where such an aerial system is used that the latter is negligible
in comparison. There is no doubt whatever that when local
interference is troublesome the use of a suitable type of anti-
interference aerial will effect a great improvement, provided
always that the aerial proper, as distinct from the lead-in, can
be erected outside the field of interference. If this cannot be
done, then the interference will still be present.
Importance of the Earth
In this connection it should not be forgotten that the earth
may be as important as the aerial. Signals and interference
can be picked up by the lead from the receiver to earth and
by the ground iUelf from the earth to a point below the aerial.
Little in the way of signals is picked up by this portion because
of its low effective height, which reaches zero in the case of
the ground. Severe interference may be picked up, however,
sometimes even more than by the aerial lead-in. The inter-
ference is often severely local and the earth lead often passes
through the most intense part of the field ; moreover, there
210
THE AERIAL-EARTH SYSTEM
Fig. 93 : This sketch shows one nnethod of erecting an aerial at a distance from a house
in order to keep it out of the interference field
may be interference currents circulating in the ground. It
must never be forgotten that the complete aerial circuit com-
prises the horizontal span of the aerial, the lead-in, the primary
circuit of the receiver, the earth lead, the conductors ('water-
pipes, soil, etc.) from the point of connection of the earth lead
to the area beneath the aerial, and the space between the
horizontal span of the aerial and the ground beneath. At any
point within this circuit interference can be picked up and
there is little to be gained by screening the down-lead if the
earthy path from receiver to the under-aerial ground passes
through the interference zone. Disappointment with anti-
interference aerials, when care has been taken to erect the
aerial itself outside the field of interference, is most probably
due to neglect of this point.
Theoretically, it is only possible to secure complete freedom
from interference when the entire aerial-earth system is removed
from the field of interference. This is not always as difficult
to arrange as it sounds, and the scheme depicted in Fig. 93 can
often be adopted. It will be seen that the aerial is erected well
away from the house, and the down-lead and earth arranged
at the end remote from the house. In normal circumstances
this will give the minimum of interference, for the whole
aerial-earth system is as far as possible from the house in which
the interference is presumed to originate.
Since it is somewhat inconvenient to keep one’s receiver at
the bottom of the garden, it is necessary to take the signals
picked up by the aerial to the receiver without interference
being picked up en route. This is done by means of the step-
down transformer and screened low-impedance cable. This
211
WIRELESS SERVICING MANUAL
latter can usually be readily disposed inconspicuously by
burying it or cleating it to the garden fence.
The difficulties are greater, of course, where no garden is
available, but even then they are not always insuperable. In
the case of a modern building such as a block of offices or
flats, the aerial must usually be erected on the flat roof and
there is usually very severe interference generated within the
building itself. If the receiver is on one of the lower stories
the conventional screened down-lead and earth may prove
satisfactory, but is unlikely to be so when the receiver is used
on an upper floor, for quite a large portion of the building is
then acting as an earth lead. The best results are then likely
to be secured by using a counterpoise earth as shown in Fig. 94.
The aerial should be erected as high as possible and a counter-
poise, consisting of several parallel wires about a foot apart
and connected at one end but otherwise insulated, erected
beneath it and used as the local earth. The counterpoise can
be about six feet above the roof. The importance of height
Fig. 94 : On a large building, a counterpoise may prove superior to an earth connection
212
THE AERIAL-EARTH SYSTEM
for the aerial will be realised when it is remembered that the
effective height will not exceed the distance between aerial and
counterpoise.
Arrangements such as these, however, are necessary only
when local interference is present. In the domestic areas of
rural and suburban districts local interference is probably
negligible in most cases, and efficiency is the chief factor to be
considered in the design of an aerial, and in the attainment of
efficiency height is the most important factor. For general
reception on all wavelengths experience shows that the ordinary
inverted-L aerial is the best. It is very difficult to improve on
this type for medium and long wave reception and in most
cases it is the best for the short waves also. An exception
occurs when reception is required on a particular wavelength
or a narrow band of wavelengths only, for then some form of
resonant aerial can be used with advantage.
The physical dimensions of resonant aerials are proportional
to the wavelength and it is consequently only possible to erect
types for the shorter wavelengths in the space available in the
normal garden. Moreover, a resonant aerial must be joined to
the receiver properly ; with most types a simple lead-in wire is
not good enough. It is necessary to use a properly terminated
transmission line, which is more usually termed a feeder, and
which if arranged correctly does not itself pick up signals.
Such aerial systems are consequently also anti-interference
aerials and a correctly arranged resonant aerial and feeder will
give as much freedom from local interference as the usual anti-
interference types. It will do this with high efficiency as long
as it is only used for reception on wavelengths close to the
resonant value ; at other wavelengths the efficiency falls off,
and the immunity from local interference is also reduced because
with the change in the aerial impedance the feeder is no longer
correctly terminated.
Use of a Resonant Aerial
In general, a resonant aerial is to be recommended when
reception in a particular narrow band is desired. Anyone who
is specially interested in lo-metrc or 20-metre amateur recep-
tion, for instance, would be well advised to erect an aerial
resonant to one of these wavelengths. Again, for some special
reason, the best possible reception of some particular station,
such as 2XAD, might be required, and here also an aerial
resonant to this wavelength would be advisable. In this
connection it should be pointed out that many resonant aerials
213
WIRELESS SERVICING MANUAL
are directional and should consequently be erected so that
best reception is obtained from the desired stations. In
working out the direction it is useless to calculate the direction
from an ordinary atlas based on Mercator’s Projection, at any
rate for distances exceeding a few hundred miles. A globe
must be used or else a map with a Great Circle Projection.
The simplest form of resonant aerial is a wire having a length
which is definitely related to wavelength. The commonest are
the A/2 and A/4 aerials (A = wavelength in metres) ; the A/4
aerial, however, is not such a true resonant aerial, for it depends
on the earth for its operation. The current distribution in such
aerials is shown in Fig. 95, and is a maximum in the centre for
the A/2 type and at one end for the A/4 type. The impedance
of the aerial is lowest when the current is greatest and vice
versa, and a half-wave aerial has a centre impedance of about
72 ohms. The end-impedance would ideally be infinite, but
in practice is of the order of 2,500 ohms.
The aerial can have the feeder connected at the end or the
centre, but the feeder must have the same impedance as the
aerial at the point of connection. Now the feeder impedance
depends upon the spacing of the two wires of which it consists
and upon their diameter, and the values obtainable are severely
limited in practice. Where the aerial and feeder impedances
are not the same a matching device must be inserted between
the two, and this often takes the form of a resonant line which
has the property of acting as an impedance transformer.
Special feeder cable with an impedance of 72 ohms is now
available, and can be connected directly to the centre of a
half-wave aerial as shown in Fig. 96 (a). The feeder should
be brought away from the aerial at right angles for a distance
Fig. 95 : The dotted lines show the current distribution in resonant aerials
214
THE AERIAL-EARTH SYSTEM
of at least a quarter-wavelength. This is easy when the aerial
is horizontal, but troublesome when it is vertical.
The choice between vertical and horizontal aerials depends
on w^hich is used at the transmitter, for the aerials at both ends
should be the same. In general, however, this is of little
importance in long-distance reception, for the plane of polarisa-
tion of the wave-front gets twisted in its passage to the receiving
aerial. It is usual, therefore, to adopt a horizontal aerial for
ordinary short-wave reception since it is more convenient to
erect. For television reception, however, a vertical aerial is
necessary, since the transmitting aerials are vertical and the
distance is normally short.
Owing to the mechanical difficulty of bringing the feeder out
at right-angles from the centre of a vertical aerial, it is usually
more convenient to feed it from the end. This necessitates
the use of an impedance transforming section for matching the
aerial to the feeder. Fortunately such a section is extremely
simple to construct for it consists merely of two parallel wires
215
WIRELESS SERVICING MANUAL
A/4 long, as shown in Fig. 96 (ft). At the lower end the two
wires of the feeder are joined to the two wires of the matching
section ; at the upper end one wire is joined to the aerial and
the other left free.
In practice, the aerial and matching section are built together
and a single length of wire is used for the aerial and one limb
of the matching section. The other limb is a length of the
same wire spaced from the first by a number of spreaders.
The spacing is important and depends on the wire diameter.
A third type of aerial is the centre-fed half-wave aerial with
a reflector. This type of aerial gives increased efficiency and
somewhat less pick-up of interference on account of its improved
directional properties. It consists simply of a A/2 aerial
centre-fed as in Fig. 96 (c) but mounted vertically. Behind it
is mounted another insulated wire of length A/2 and spaced
from the aerial by A/4. In this case it is quite easy to bring
out the feeder at right-angles to the aerial, for it can be supported
by the framework which is needed for holding the reflector.
This type of aerial is widely used for television reception.
The actual lengths of resonant aerials are always slightly
shorter than the nominal lengths by which the types are known,
for the diameter of the conductor and the method of supporting
it affect the resonance frequency. 'I'he length of a A/2 aerial is
thus not exactly one-half wavelength, but must be about 0*94
A/2. Full details regarding the calculation of aerial lengths and
feeders are given in Appendix I.
It is worthy of note that resonant aerials of the A/2 type do
not depend upon earth to complete the circuit but arc complete
in themselves. The earlier remarks regarding the importance
of the earth in interference elimination thus do not apply to
these types. They are actually of an ideal form for the avoidance
of local interference, but the aerial itself must naturally be
erected outside its field of influence. Resonant aerials should
always be erected as high as possible, for not only does the
field strength rapidly increase with height at ultra-short wave-
lengths, but ignition interference from cars rapidly decreases.
216
CHAPTER 22
AUTOMATIC FREQUENCY CONTROL
Q uite a number of set makers have in the last few years
produced receivers which include some form of auto-
matic frequency control, or A.F.C. for short. (It is also
sometimes less correctly known as automatic tuning control,
or A.T.C.). It is, however, by no means common as yet, and
because of its unfamiliarity its adjustment may prove somewhat
troublesome and it may even complicate the ordinary processes
of fault-finding. It is, therefore, desirable to treat it in rather
great detail.
It is first necessary to be perfectly clear about what A.F.C.
is and how it functions. The purpose of A.F.C. is to make it
impossible for a receiver to be appreciably mistuned from a
signal. In the case of a selective broadcast receiver quite a
small amount of mistuning has a marked effect upon the quality
of reproduction. In spite of this, many people seem incapable
of tuning a receiver correctly and habitually leave it mistuned
by amounts up to several kilocycles a second.
When A.F.C. is fitted it suffices if the tuning control is set
within some 5 kc/s of the frequency of the desired signal. The
set then tunes itself to the signal with quite a high degree of
accuracy. It is obviously of great value in motor-driven push-
button sets and is frequently included in them.
A.F.C. is applied only to the superheterodyne and it functions
by varying the oscillator frequency so that the difference
between the signal and oscillator frequencies is brought exceed-
ingly close to the intermediate frequency. The signal-frequency
circuits are not affected, so that if the tuning is not adjusted
close to resonance by the manual control in the first place, the
result is to leave the signal-frequency circuits mistuned. As a
rule this is not very important because they are usually fairly
flatly tuned.
An A.F.C. system consists of two main parts — the frequency-
control stage and the discriminator. The former is a circuit
arrangement which behaves as a reactance, the value of which
depends upon the magnitude and polarity of an applied voltage,
while the latter is a device for producing a voltage of value
dependent upon the amount of mistuning and polarity
dependent upon the direction of mistuning.
The frequency-control stage almost invariably employs a
valve ao connected that it behaves as a reactance, the value of
in
11
WIRELESS SERVICING MANUAL
(a) (b)
Fif. 97 : Four frequency-control circuits are shown here ; (a) and (d) are
equivalent to a variable capacity and (b) and (c) to a variable inductance
which is controllable by varying the grid bias. There are many
possible circuits and the chief ones are shown in Fig. 97.
Consider the circuit (a) and assume that the reactance of C
at the operating frequency is very largely compared with the
resistance R. Then the current through R and C leads the
applied voltage by very nearly 90^" and the grid voltage leads
the applied voltage by the same amount, since it is developed
by the passage of this current through R. The anode current
of the valve is in phase with the grid voltage, and consequently
leads the applied voltage by nearly 90°. A leading current,
however, is that drawn by a condenser, consequently the circuit
of Fig. 97 (a) behaves as a condenser, the value of which
depends on the alternating anode current. This in turn depends
218
AUTOMATIC FREQUENCY CONTROL
on the mutual conductance of the valve, which can be altered
by changing the grid bias. We thus find the circuit to be
equivalent to a variable condenser. By similar reasoning, Fig.
97 (6) can be shown to be equivalent to a variable inductance.
In this circuit Ci is merely a blocking condenser and serves
only to insulate the grid circuit from the H.T. supply.
An Electronic Inductance
The arrangement (c) is somewhat different. I'he basic com-
ponents are R and C ; Ci is only a blocking condenser and Ri
provides grid circuit continuity. In this case assume that the
reactance of C is very small compared with the value of R.
Then the current through R and C is nearly in phase with the
applied voltage and the grid voltage, developed across C, is 90°
lagging on the applied voltage. The anode current is in phase
with the grid voltage and is consequently 90° lagging on the
applied voltage. The circuit as a whole thus behaves as a
variable inductance. By similar reasoning (d) can be shown to
be equivalent to a variable condenser.
If the input terminals of any of these circuits are connected
across the tuned circuit of the oscillator in the frequency-
changer it is clear that the oscillator frequency can be varied
by altering the grid bias of the control valve.
It should, of course, be understood that the circuits shown
in Fig. 97 are not complete, but merely the basic arrangements.
Also, the input impedance is not a pure reactance but always
contains a resistive component which damps the oscillator
circuit somewhat.
Any of these basic circuits can be used for frequency control,
but both (b) and (d) are rarely found in practice. The use of
an inductance for the reactive element in the circuit has two
disadvantages ; it must be screened to restrict its magnetic
field and the self-capacity of the coil is liable to modify the
action somewhat.
Of the two remaining circuits (c) is the one commonly used.
It tends to have a higher input resistance than (a), which is its first
point of advantage. Of greater importance, however, is the fact
that it behaves as an inductance whereas (a) behaves as a
capacity.
The use of a variable inductance effect is advisable because
it only necessitates some increase of the oscillator coil inductance
to give correct ganging. A variable capacity effect, on the
other hand, means that the oscillator circuit capacity must be
reduced and this is not always possible.
219
WIRELESS SERVICING MANUAL
This must be clearly understood if the action of the circuit
is to be followed. Take the arrangement (a) acting as a
variable capacity. It gives a range of capacity from Cmin. to
Cmax. If the receiver is mistuned so that the oscillator
frequency is too high the bias on the control valve is reduced
to increase its mutual conductance and so increase its input
capacity. On the other hand, if the receiver is mistuned the
other way, so that the oscillator frequency is too low, the bias
is increased and the mutual conductance and input capacity
fall. In both cases the oscillator frequency is brought nearer
to its correct value.
Now it is clear that the receiver must be initially ganged
with the control valve giving an input capacity of about the
mid-point of its range, for otherwise it would not be possible
to obtain an equal increase or decrease of capacity for tuning
correction. The ganging must be carried out with the control
Cm 1- C
valve adding a capacity of about ~ to the circuit.
Now this means that the normal oscillator circuit stray
capacity must be reduced by the same amount to enable correct
ganging to be secured. The circuit capacity is normally kept
as small as possible, however, in order to secure a wide tuning
range, and it is consequently often impossible to reduce it
further by the requisite amount.
With the inductive input circuit (c) no change of oscillator
circuit capacity is needed beyond a small reduction to com-
pensate for the wiring capacity and the anode-cathode capacity
of the control valve. This is relatively small and can usually
be accomplished without undue difficulty.
The control valve has an inductance varying from Lmin. to
Lmax. with a mean value of It is only necessary,
therefore, to increase the oscillator inductance so that when it is
in parallel with this value it has its original value. If this last
is L, the new value Ll must be (Lmln. h Lmar.) / (L,ntn. \- Lmax.
— 2 L). This can always be done.
A typical control stage of this type is shown in Fig. 98 con-
nected to a frequency-changer of the triode-hexode type. The
oscillator circuit is quite conventional and the control circuit
differs from that of Fig. 97 (c) only by the inclusion of the
various feed components.
Ci and C2 are both blocking condensers and are necessary
only to maintain the correct D.C. paths. They are of fairly
large capacity, say o-ooi fiF., but are in no way critical. Ra is
220
AUTOMATIC FREQUENCY CONTROL
the resistance through which the control voltage is applied and
is again not critical as long as it is large compared with the
reactance of C3. R4 and C5 are the usual screen-feed com-
ponents and R3 and C4 are for cathode bias. R3 is chosen to
give a standing bias towards the middle of the valve charac-
teristic. It must not be too small, for the A.F.C. control
voltage may be either positive or negative with respect to the
earth line. The important components are Ri and C3 for they
control the value of the input inductance of the stage.
The Discriminator
Now it will be clear that in order to effect the control of the
oscillator frequency it is necessary to apply to the control valve
a bias voltage the value of which depends upon how much the
oscillator frequency differs from the correct value and the
polarity of which depends upon whether its freciuency is higher
or lower than the correct value.
Since the intermediate frequency is dependent on the
oscillator frequency this voltage can conveniently be derived
from it and it is done by means of a circuit known as the dis-
criminator. I'here are two main types of circuit — the older is
much the simpler to understand, but is much more difficult to
adjust. Consequently, it is not often used now. One form
of this older circuit is shown in Fig. 99. Vi is the last I.F.
Fl«* 98 : This diagram shows a typical arrangement of a frequency-control stage
with the oscillator of the frequency-changer
221
WIRELESS SERVICING MANUAL
stage and the circuit LiCi is tuned to the desired intermediate
frequency. It is fed to the diode V2 through C4 ; this diode
acts as a detector and A.V.C. source — the A.F. signal being fed
out in the usual way through C7 and the A.V.C. voltage being
taken off through the filter R6 C9.
All this is quite conventional. We now come to the A.F.C.
circuits. These consist of the two diodes V3 and V4 fed from
L2 C2 and L3 C3 and with load circuits R3 C5 and R4 C6 respec-
tively. L2 and L3 are each coupled to Li by the same amount
so that equal voltages are induced in each.
If these two circuits are tuned to the same frequency the
I.F. voltages applied to V3 and V4 will always be equal, and if
the load circuits are identical the output of each diode will be
the same. It will be noted that the cathode of V3 is joined to
earth and the A.F.C. voltage is taken off through R5 C8 from
the cathode of V4. The anode ends of the two load circuits are
joined together.
The total output is thus the difference between the voltages
222
AUTOMATIC FREQUENCY CONTROL
across R3 and R4. When they are equal the resultant is zero.
Under the conditions given above — Lz Cz and L3 C3 tuned to
the same frequency — both diodes give equal outputs and the
resultant on the A.F.C. line is always zero.
In practice these circuits are not tuned to the same frequency.
One is tuned above the intermediate frequency and the other
below.
Both circuits must be carefully tuned precisely to the right
frequencies. Using the control circuit of Fig. 97 (c) we have
seen that if the oscillator is at too high a frequency we need an
increase of inductance to correct it and that this necessitates a
reduction of grid bias on the control valve.
This in turn means that the A.F.C. voltage must be positive
so that the output of V4 in Fig. 99 must predominate over that
of V3. Now when the oscillator frequency is too high it means
that the intermediate frequency is too high also — in the normal
case when the oscillator frequency is higher than that of the
signal — consequently, L3 C3 must be tuned above the inter-
mediate frequency.
What happens is this — when the intermediate frequency is
higher than it should be a greater voltage is set up across C3
than across Cz because L3 C3 is nearer resonance with it than
is Lz Cz. The output of V4 exceeds that of V3 and the A.F.C.
voltage is the difference of the two, and positive.
This operates on the control valve to lower the oscillator
frequency and reduce the intermediate frequency also. As the
intermediate frequency comes nearer to the correct value, the
voltage on V4 decreases and on V3 increases. Their outputs
vary similarly and the. difference decreases. An equilibrium
condition is soon reached at which the amount of detuning is
very small.
There must, of course, always be some degree of mistuning,
otherwise no control voltage could appear. If the intermediate
frequency is exactly right, the outputs of the two diodes
are equal and the A.F.C. voltage is zero. A.F.C. can thus
never produce perfect tuning ; what it does is to make it
impossible for the tuning errors to exceed a very small amount
fixed by the designer and which is usually a few hundred cycles
at most.
Initial Adjustments
When adjusting a system of this nature it is advisable to
follow carefully the instructions appertaining to the particular
receiver if they are available. If they are not, the following
procedure will usually be satisfactory.
223
WIRELESS SERVICING MANUAL
Referring to Fig. 99, set C3 to its minimum capacity and C2
to its maximum, and short-circuit the discriminator output.
This last is conveniently done by short-circuiting C8. Then
adjust the I.F. circuits, including Li Ci, and the ganging
of the signal and oscillator circuits, in the usual way.
When satisfied that the performance of the set without A.F.C.
is in every way correct it is time, and then only time, to adjust
the A.F.C. circuits. If the frequencies to which the dis-
criminator circuits should be tuned are known the adjustment
is not difficult.
Connect a test oscillator to the I.F. amplifier in the same way
as for I.F. alignment and a low-range milliammeter in series
with R4. Set the test oscillator to the higher of the two A.F.C.
circuit frequencies and adjust C3 for maximum output.
Then set the test oscillator to the lower of the two frequencies
and adjust C2 for maximum output with the milliammeter
joined in series with R3.
These adjustments may not be sufficiently precise for good
results, since it is often impossible to set the test oscillator to
the desired frequencies with sufficient accuracy. The next
step, therefore, is to set the test oscillator to the intermediate
frequency and connect a valve voltmeter across C8. The valve
voltmeter must be capable of responding to a steady potential,
so it must not have a condenser input circuit.
If the circuits are correctly adjusted no reading should be
found on the voltmeter under these conditions and on detuning
the test oscillator the voltage should rise with one polarity for
mistuning one way and with the other polarity for mistuning
the other. The magnitudes of the voltages for equal amounts
of mistuning should be nearly the same.
It will probably be found that with the test oscillator at the
intermediate frequency zero output is not obtained. C2 or C3
should then be adjusted to bring the output to zero. If equal
outputs for equal degrees of mistuning are not then obtained
the cause is probably unequal coupling between Li and L2 on
the one hand and Li and L3 on the other, or else unequal
effective Q values for the tuned circuits.
A.F.C. Circuit Frequencies
If the alignment frequencies for the A.F.C. circuits are
unknown suitable ones will have to be found by trial. For a
start it is satisfactory to take them as 5 kc/s above and below
the intermediate frequency, and to adjust in the manner just
described.
224
AUTOMATIC FREQUENCY CONTROL
Fig. too : With a modern discriminator both tuned circuits are tuned to the
intermediate frequency and the action depends on the phase difference of the
primary and secondary voltages
If the frequencies related are too near the intermediate
frequency the band over which the A.F.C. operates will be
too small. In other words, A.F.C. will be capable of correcting
for only a small degree of mistuning.
On the other hand, if the frequencies are too far from the
intermediate frequency, A.F.C. will operate over too wide a
band. If this happens, A.F.C. will prove most unsatisfactory,
for it will be liable to be affected by signals on the adjacent
channels to the wanted station.
It may then prove impossible to receive a weak signal with
a fairly strong one on the next channel, for the set will be liable
to tune itself to the stronger. This will be found particularly
distressing when there are two fading signals on adjacent
channels, for it is quite possible for the set to keep tuning itself
from one to the other, keeping in tune with whichever fading
makes the stronger for the moment.
The remedy for this state of affairs is to avoid too great an
amount of detuning of the A.F.C. circuits and to keep the
I.F. selectivity fairly high.
Largely owing to the difficulty of adjustment this particular
A.F.C. system is not much used nowadays. A more common
arrangement is one similar to that shown in Fig. loo. Here
both I.F. circuits are tuned to the intermediate frequency, so
that the adjustment is but little different from that of any
ordinary amplifier.
225
WIRELESS SERVICING MANUAL
The circuit shows that there are, as before, two diodes Vz
and V3 following the last l.F. stage Vi. The diodes have the
load resistances Ri and Rz so connected that the outputs oppose
one another on the A.F.C. line, which is taken through the
filter R3 C5.
The essential difference between the circuits lies in the
method of obtaining the inputs to the diodes. Instead of using
separate input circuits tuned to different frequencies, common
circuits tuned to the intermediate frequency are adopted.
Phase-Variation System
The action depends upon the phase difference between the
currents in Li and Lz. The secondary Lz is centre-tapped
and one-half of the voltage across it is applied to one diode while
the other half is applied to the other diode. These two voltages
are 180° out of phase with each other relative to the centre
point.
The voltage developed across the primary is also applied to
the diodes via C3 connected to the centre tap on the secondary.
The total voltage applied to one diode is thus the primary
voltage plus one-half the secondary voltage and that applied
to the other diode is the primary voltage, minus one-half the
secondary voltage. The addition and subtraction must be per-
formed vectorially, however.
Now when both circuits are tuned to the same frequency
the secondary current is 90^ out of phase with the primary
current at resonance. The voltages developed across primary
and secondary are consequently also 90° out of phase. The
input to one diode is thus the primary voltage plus a voltage
leading it by 90° and to the other diode is the primary voltage
plus a voltage lagging on it by 90°. These voltages add vectori-
ally to equal magnitudes, so at resonance the diodes have equal
inputs. The outputs are thus also equal and add to zero.
Away from resonance the primary and secondary voltages are
still equal for the same degree of mistuning from resonance, but
their relative phase angles are different. The voltage induced
in the secondary is still 90° out of phase with the primary
current, but away from resonance the secondary current is no
longer in phase with this induced E.M.F.
Suppose the degree of mistuning is such that the secondary
cur/ent is 30° out of phase with the induced E.M.F. , Then the
voltage applied to one diode consists of the prin>ary voltage
plus one-half secondary voltage at 60° to it, and to the other
diode it is the primary voltage plus one-half secondary voltage
226
AUTOMATIC FRE(iUENCY CONTROL
at 120° to it. The two add up to different magnitudes and the
diode inputs are consequently unequal.
The band of frequencies over which control is secured is
governed by the Q of the tuned circuits and the coupling
between them. The user, however, is less concerned with
these than with the adjustment of the circuits. They must be
adjusted for resonance at the intermediate frequency, and to a first
approximation this may be done by trimming them for maximum
output as indicated by a meter connected in series with Ri.
With over-coupled circuits, how'ever, this alone is not likely
to be sufficient and it will genexally be necessary to check the
adjustment. The first step is to make sure that the diode inputs
are equal at the intermediate frequency with the aid of a valve
voltmeter across the A.F.C. line.
Having obtained a zero voltage across this line, check that equal
degrees of mistuningon either sideof resonance give approximately
equal magnitude of A.F.C. voltage but of opposite polarity.
Faults which may develop in A.F.C. systems are of the same
nature as those in other parts of the receiver and can be traced
by normal testing methods. In general fault-finding on an
A.F.C. -equipped receiver, of course, it is a wise plan to dis-
connect, or otherwise render inoperative, the frequency-control
stage. The simplest and most satisfactory way is usually to
short-circuit the A.F.C. line.
Fault-finding in the other parts of the receiver can then be
done on the normal lines dealt with in the earlier parts of this
book. Troubles which may develop in the A.F.C. system itself
are usually of a fairly simple nature and probably the commonest
is misalignment of its tuned circuits. This can have a very
serious effect on the performance of the receiver, for instead of
A.F.C. performing its proper function of keeping the receiver
always tuned to the signal it will then consistently keep it mis-
tuned from the signal.
When an A.F.C. -equipped receiver behaves as though it were
always somewhat mistuned from a signal — and this is evident
by its effect upon the quality of reproduction — it is almost
certain that the trouble lies in the adjustment of the tuned
circuits in the discriminator.
Other troubles, such as unequal outputs from the diodes
and short-circuits, partial or complete, on the A.F.C. line are
most easily detected with the aid of a D.C. valve voltmeter. An
ordinary voltmeter is not of great assistance, for its resistance
will be low compared with that of the circuit and its indications
will consequently be misleading.
227
CHAPTER 23
PUSH-PULL AMPLIFIERS
M uch confusion exists in regard to push-pull amplifiers,
and defects are often suspected when none in fact exists.
A common complaint is that the removal of one of the
push-pull output valves causes a very srpall drop in volume
and does not apparently introduce distortion. This is no
indication of any defect and is an entirely nonnal result. It
must be remembered that although the removal of one valve
halves the output, this represents a drop in volume of only
3 db. which is by no means a large amount. Furthermore,
distortion is not always apparent because the removal of one
valve lightens the load on the mains equipment with the result
that an abnormally high voltage is applied to the remaining
one and its undistorted output is consequently increased.
Because of this rise in voltage, the practice of withdrawing
one output valve must be considered unwise, and it is definitely
dangerous to valve life to do so when both output valves have
a common bias resistance. The remaining valve will then be
grossly overrun and is unlikely to survive such treatment
unharmed.
Another effect often found on removing one valve is that
whereas the removal of one has little effect, the removal of the
other instead results in motor-boating. Again this is no indica-
tion of a defect. One of the great advantages of push-pull is
that less decoupling is needed for the avoidance of feed-back
effects than if a straight amplifier were used. All the advantage
of the push-pull connection is lost when one output valve is
removed and it is consequently only natural that feed-back
effects should manifest themselves. Since the phase is opposite
on the two sides of the push-pull stage, it follows that on the
removal of one valve the feed-back will be degenerative and
have little audible effect^ whereas on the removal of the other
it will be regenerative and may cause motor-boating.
True defects in a push-pull stage are usually of a fairly
simple nature and consist of widely dissimilar valves and
defective input or output transformers which can be checked
by normal methods.
There are, however, some important differences between
transformer and resistance coupled amplifiers. In the form, in
which is it usually found the transformer-coupled amplifier is
balanced in each stage individually, and it is balanced not only
for the signal but for feed-back. The resistance-coupled
228
PUSH-PULL AMPLIFIERS
amplifier does not provide an individual stage balance, but a
balance on the signal for the complete amplifier only, and in
most cases it is not balanced against feed-back.
A typical transformer-coupled amplifier is shown in Fig. loi.
The first push-pull stage consists of V2 and V3 and is fed from
an unbalanced input stage Vi. The output stage V4, V5 is
again push-pull. The main advantage of push-pull lies in the
output stage, for if the valves are alike even-order harmonics
distortion is balanced out, and only odd-order harmonics
appear in the output. With triodes, second harmonic distortion
becomes apparent well before third harmonic in individual
valves, so that the output obtainable from two valves in push-
pull is greater than that from the same two valves in parallel.
In addition to this, the anode current flows in opposite
directions through the two half-primary windings of the output
transformer T3, and its core is consequently relieved of the
magnetization inherent with a single-ended stage. This is
very important, for it is economically almost impossible to
design a transformer with a wide frequency response and a
low distortion level for a large output stage of the single-ended
type, whereas it is relatively easy with push-pull.
A transformer-coupled push-pull output stage is usually fed
from an unbalanced stage, and a push-pull penultimate stage
like that of Fig. 10 1 is used only when a single valve will not
give sufficient output to feed the output stage. In general, a
double push-pull amplifier is used only when the output valves
need a very large signal input — of the order of 100 volts per
valve or more.
The most likely defect in a transformer-coupled push-pull
stage is a serious disparity between the valves. It is not
difficult to check the balance of a stage. The easiest way is to
connect a pair of headphones, shunted by a loo-ohm resistance,
in series with the lead to the output transformer centre-tap.
Disconnect the centre-tap on the input transformer and connect
across the secondary a potentiometer with its slider joined to
the bias source. The potentiometer can well have a resistance
of the order of o’2 megohm.
Varying the setting of this new and temporary control alters
the ratio of the inputs to the two valves. The procedure,
therefore, is to apply an input to the amplifier, preferably at
about 400-1,000 c/s, and adjust the potentiometer for zero
output in the phones. If the input is too great a good zero
will not be obtained, but there will be a balance point for the
fundamental and the second harmonic alone will be audible.
229
WIRELESS SERVICING MANUAL
Now check the resistance between the slider of the potentio-
meter and each of its ends, disconnecting it from the transformer
while doing so. If the two values are the same, or nearly so,
the valves and output transformer are sufficiently well balanced.
If they are markedly different, however, interchange the two
valves, rebalance, and measure the two new resistance values.
If they are the same as before, the valves are alike, and the
unbalance is in the output transformer. If the resistance
values are also inter-changed, however, the output transformer
is in order but the valves are unlike. Of course, if the resistance
values have neither stayed unchanged nor inter-changed, both
valves and transformer are at fault.
The input transformer is less easy to check, and the best
way is probably to measure the voltages across its two halves
with a valve voltmeter. A rough check can be obtained, how-
ever, during the preceding tests with the potentiometer. Adjust
the input level so that a fairly sharp balance on the control is
obtained, then change over to the normal input connections
without the potentiometer, and note whether the fundamental
is still inaudible in the phones. If it is, the input transformer
is probably satisfactory.
It will be noted that the bias resistances R3 and R4 are
shown in Fig, loi without by-pass condensers. This is good
practice when the stages are Class “ A for the resistance
gives a considerable degree of self-balancing action to the
stage and makes inequalities in the valves of much less
importance. It does tend, however, to increase the odd-order
harmonic distortion in the output, but the increase is quite
negligible for degrees of second harmonic distortion per valve
up to some 10 per cent.
The omission of a by-pass condenser is permissible in all
Class “ A ” stages and also in Class “ AB *’ stages in which the
non-linearity per valve is not large. It is not permissible in
Class “ B ” stages, nor in Class “ AB ” stages having large
non-linearity per valve. As a rough guide if on applying a
signal giving full output the anode current for the stage rises
above the value with no signal by more than about 10-15 per
cent, a bias resistance by-pass condenser should be used.
It will be noted from Fig. 10 1 that each push-pull stage is
balanced independently of the other. The input division and
output combination is done in the transformers. Even the
removal of a valve from the penultimate stage disturbs the
balance of the output stage in no way.
The amplifier, too, is completely balanced as far as feed-back
230
PUSH-PULL AMPLIFIERS
Fig. 101 : A two-stage transformer-coupled push-pull amplifier takes the
form shown in this diagram. The circuit is rarely used except in very large
power amplifiers
effects from a common H.T. supply impedance are concerned.
Imagine each push-pull stage to be perfectly balanced. Then
in either stage any disturbance on the H.T. line will cause
equal changes of anode current in the valves of a stage, and
these will cancel in the transformer and induce no voltage in
the secondary.
A disturbance in the H.T. supply to the first valve Vi,
however, will pass right through the amplifier, for it will act
like an input signal and induce equal and opposite voltages in
the two halves of the secondary. The anode currents of the
following valves will change by equal and opposite amounts,
so that there will be no change in the total current drawn by
a push-pull stage.
Although a disturbance in the H.T. supply to the first
valve may pass through the amplifier, because it does not
affect the combined anode currents of the push-pull stages, it
cannot produce a fresh disturbance on the H.T. line. Feed-back
will not exist, and there is no need to decouple the first stage.
In practice, with a two-stage push-pull amplifier, it is usually
wise to include some decoupling, because no practical amplifier
ever is perfectly balanced. Moreover, distortion may occur at
231
WIRELESS SERVICING MANUAL
large outputs through harmonics of the signal being fed-back.
This is more likely with Class “ AB than with Class “ A
With only a single push-pull stage, it is rarely necessary to
decouple the penultimate stage, but earlier stages must be
decoupled.
Transformer coupling in push-pull stages is used chiefly
with Class “ AB2 *’ and Class “ B^ ” output stages, and resistance
coupling is more common with Class “ A and Class “ AB^ ”
stages. A resistance-coupled push-pull amplifier comprises
four items — a phase-splitting input device, which may be a
transformer or some valve arrangement, two identical R.C.
amplifiers, and an output pha^e-combining device, which is
Fig, t0!2 : A two-scae« rctistance-couplttd push-puli amplifier is frequently
found in high quality equipment. In general, the bias resistances do not need
by-pass condensers
232
PUSH-PULL AMPLIFIERS
invariably a t r a n s -
former. In common
parlance, the output
stage consists of the last
valve in each amplifier
taken together, and the
penultimate stage con-
sists of the last but one
in each amplifier. Thus,
in Fig. 102, Vi and V3
form one of the R.C.
amplifiers and V2 and V4 the other, but V3 and V4 form the
output stage, and Vi and V2 the penultimate stage.
This diagram contains only three of the four items mentioned
above, for the input circuit is not shown. This is because
there are so many different types, and a selection of them is
illustrated later in separate diagrams. Referring again to
Fig. 102, the important thing to note is that the two stages are
not independently balanced as in the case of transformer
coupling. If V3 and V4 are identical but Vi and Vz are not,
the whole amplifier is unbalanced and not merely the penul-
timate stage. Save for the bias resistances, the two amplifiers
are completely independent between the input and output
circuits.
If the individual sides of a stage are alike, and its valves have
equal inputs in opposite phase, the alternating currents through
the bias resistance are equal and opposite. Feed-back effects
are then non-existent. If the sides of a stage are not identical,
however, the bias resistance provides a self-balancing action
just as in the case of transformer coupling. The omission of a
by-pass condenser is equally, and under the same conditions,
good practice.
In order to secure good balance it is not sufficient for the
valves and their inputs to be alike ; the coupling resistances
must also be matched. In Fig. 102, for instance, Ri and R2
should have the same value, while R3 should equal R4. A
reasonable degree of equality would be within 5 per cent., but
2 per cent, is better, especially if there is no by-pass condenser
across R5. This is because feed-back on R5 accentuates
unbalance caused by inequality of the coupling resistances,
while reducing that produced by the valves. It is easier to
match resistances than valves, so that the condenser is usually
omitted.
The equality of the couplings is easily checked by n^easuring
233
fig. 103 : As explained in the text, the balance
of a push-pull ampHrier can be checked with
this simple apparatus
WIRELESS SERVICING MANUAL
the resistance and capacity values. The first stage balance can
then be checked by applying an input from an A.F. oscillator
through the circuit of Fig. 103 and listening with a pair of
headphones connected across R5. A condenser should be
connected in series with the phones to avoid disturbing the bias.
The potentiometer is adjusted for zero fundamental com-
ponent in the phones and the two halves of its resistance
checked, exactly as described for transformer-coupled ampli-
fier. When good balance has been obtained in the first
stage, the second can be tackled in the same way, but this time
listening with the phones connected across a low resistance in
the lead to the output transformer centre- tap.
The phase-splitting arrangements for resistance-coupled
push-pull amplifiers vary greatly, and some of the commonest
arrangements are shown in Figs. 104-108. The first is that of
Fig. 104 and it will be seen that it consists of a direct connection
between the input and one side of the output, and a valve
stage between the input and the other side of the output.
There is a reversal of
phase in this valve
stage, and the correct
output is secured when
it gives unity amplifica-
tion. The output is
taken from the junction
of Ri and R2, which
must be proportioned
correctly. The balance
condition is /x = i +
(Ra -f Ri) (R2 + R3)/
R2R3. Typical circuit
values might be /x = 40 ;
Ra = 15,000 n ; Ri =
40,000 il ; R3 = 0*25
M Q. ; and Rz = i ,400 .
A modification of the
circuit is shown in Fig.
105. This has the
advantage of giving a
slightly lower hum
level and is more
frequently used, par-
Fif. 104 : Phase>reversai for one push-pull ticularly in American
input is obtained by an R.C. suge of unity gets. The balance
amplification
234
PUSH-PULL AMPLIFIERS
condition is (i + ^ ■ — . ■
Ra/Ri) (i -}■ R3/R4) “K . + HT
Ra/R4. Typical values <
are At= 40 ; Ra =
15.000 ; Ri = f
50.000 Q ; R3 = 0 25 t
MO : and R4 = r 5 -.r
8,800 O. ^’1
Another common , — ...
circuit is shown in Fig. Xr
106 (a). The two out- ^ I A
puts are taken from the | I ^
anode and cathode 1/ —
circuits and are conse- I
quently in opposite I — ■ — o
phase. They are equal — >
in value if R3 = Ri -f 11 >R
R2 and R5 = R6. Bias s -r >
is provided by R2 which I 1 [ , ^ | q
usually has a value of '
some 2,000 ohms, while
Ri and R3 may be
48.000 ohms and 50,000
ohms respectively. ^
Sometimes C3 is joined * ■■ ■■■ — ■■
directly to the junction Fig. 105 : This diagram is a modification
nf Rt and R2 and some- output being taken
times R 2 IS shunted by tapped coupling resistance
a large capacity ; in
both these cases R2 does not contribute to the output volt-
age and so the balance condition becomes Ri = R3. The
circuit of Fig. 106 (b) is identical save for the method of
obtaining grid bias. Instead of deriving it from the voltage
drop across a resistance in the cathode circuit, it is taken
from a tapping on a potentiometer R4, R5 across the H.T.
supply. This variation of the circuit is not often found.
The main practical difficulty that may be encountered with
this circuit is hum arising from poor heater-cathode insulation
in the valve. There is usually little difficulty if one side of the
valve heater is earthed, but there will undoubtedly be trouble
if the heater is joined to cathode or left floating. Earthing the
heater means that the cathode voltage appears across the
heater-cathode insulation, but as it rarely exceeds 50 volts,
most valves will withstand it without difficulty.
WIRELESS SERVICING MANUAL
With some people this circuit has acquired a bad reputation
for hum, but the author has used it extensively without any
difficulty at all, and he considers it the most reliable and
satisfactory of all the phase-splitting circuits. In one amplifier
embodying it, the valve has had some 10,000 hours operation
in the circuit and it still functions without hum.
It should be pointed out that the grid circuit of the valve is
of unusually high impedance and so unusually sensitive to
pick-up from the electric hum-field. The grid leak R7 is
usually about 2 megohms, but to any voltage applied between
grid and earth its value is much more than this. The cathode
voltage changes are nearly equal to the grid voltage changes,
both with respect to earth, and only the difference between
these voltages can drive alternating current through the grid
leak. The effective value of the grid leak is some 15-20 times
its actual value, and may well be 30-40 megohms. Because of
this it is possible to make C4 about one-twentieth of the normal
value without a loss of bass response. If this is done, however,
the effective impedance between the grid of the valve and
PUSH-PULL AMPLIFIERS
earth will be very high
indeed, and the liability
of the stage to hum pick-
up will be about twenty
times as great as that
of a normal stage. If
C4 is given a normal
value, however, the
impedance at hum fre-
quencies will be normal,
and the stage will not
be unduly sensitive to
hum pick-up. Never-
theless, it is well to keep
the leads between the
grid and C4 and R7
very short, and well
away from any mains
equipment.
Another phase-split-
ting circuit occasionally
employed is that of Fig.
107. One output is taken
from the anode of Vi and
is in opposite phase to
the input. The voltage
across the cathode re-
sistances R3 and R4 is
applied to the cathode of
V2 , this voltage is in the pjg ; in this phase-splitter V I acts as a cathode*
same phase as the input follower to feed a cathode- input stage V2, as well
and produces an output “ p™''*'**"* •"
at the anode of V2 which is still in the same phase.
If Ri and R2 are equal, equal outputs demand equal
alternating anode currents, but as the currents are in opposite
phase, this would mean that there could be no alternating volt-
age across R3 and R4. There would then be no input to Vz,
and there could be no output from it. The alternating
currents in the two valves cannot, therefore, be equal.
The condition for equal outputs is that ^ = i +
_ Ra2 R2
(R3 + R4) (l + f*2)
where /i2 and Ra2 are the amplification
WIRELESS SERVICING MANUAL
factor and A.C. resistance respectively of V2. The current
unbalance is then (R2 — Ri)/(R2 + Ri). The current un-
balance should preferably be less than 5 per cent., which means
R2 = I -iRi and this in turn means R3 + R4 =10
Typical values are /ai = ft2 — 40 ; Rai = Ra2 = 15,000
ohms ; Ri = 50,000 ohms ; R2 = 55,000 ohms ; R3 R4 =
17,000 ohms. For correct bias, R3 would be about 1,000
ohms, making R4 some 16,000 ohms.
One disadvantage of this circuit is the high positive cathode
potential, for the steady anode currents of both valves flow
through R3 and R4. This particular phase-splitter is usually
employed to feed an output stage directly, and then a current
of 2 mA. or more per valve is needed. This makes the cathode
potential 70 volts or more above earth. With some valves this
is an excessive diflFerence of heater-cathode potential, and yet
the heater must be effectively earthed if hum is to be avoided.
The difficulty can be got over by using a separate heater winding
for Vi and V2 ; it can then be connected to earth through a
large capacity condenser and to cathode through a high
resistance. The heaters are then earthed for A.C., but con-
nected to cathode for D.C. This same method of arranging
the heater circuit can also be employed with the phase-splitter
of Fig. 106. It is usually unnecessary, however, for as this
valve usually precedes the penultimate stage it is possible to
secure sufficient output without an excessive heater-cathode
potential difference.
The next circuit is the paraphase arrangement of Fig. 108.
Again this is usually employed immediately before the output
stage, and it will be seen to consist of two R.C. stages in which
one is fed from the junction of two resistances joining their
anodes. Its advantage is that some self-balancing action is
secured. The grid potential of V2 depends on R6 and R7,
but also on the anode potentials of Vi and V2. If the gain of
V2 falls for some reason the output of V2 falls, and the current
through R6 and R7 is less. Consequently the voltage drop
across R6 is reduced, and the grid potential of V2 increased.
This tends to increase the input of V2, and the result is that
the fall in output of V2 is not equal to the drop in the gain of
this valve.
When the two valves are alike and when R4 = R5 and
R8 = R9 the balance condition is given by
238
PUSH-PULL AMPLIFIERS
where Rii
R 4 R 8
R4 “ 1 “ R 8
Rs Rq
R5 + Rq
Typical conditions are /i = 40 ; Ra = 15,000 O ; Rz =
O'sMri ; R3 == 1,000 LI ; R4 = R5 = 50,000 ; R6 —
87,500 n ; R7 = 100,000 ; R8 = R9 =0*25 MQ.
It will be noted that in all these circuits H.T. decoupling is
shown. This is essential, because none of them is balanced
from the viewpoint of the H.T. line, and this is a disadvantage
of R.C. push-pull as compared with transformer coupling.
As an example, imagine the phase-splitter of Fig. 106 (a) con-
nected in front of the amplifier of Fig. 102. A disturbance on
the H.T. line — say, a rise of voltage — increases the anode current
of the phase-splitter. Let the rise of voltage be E and the
current L Then if R3 = Ri -\- Rz (Fig. 106 (a)) the
output at the upper terminal is E — iR^ and at the lower
terminal it is iR^. These two outputs are in the same phase
and unequal.
The grids of Vi and Vz of Fig. 102 are thus driven positively
by unequal amounts, and consequently the grids of V3 and V4
are driven negatively by unequal amounts. The anode currents
of V3 and V4 fall and as they both change in the same direction
they are additive in the H.T. lead. The total anode current of
the stage thus falls and also the drop across the impedance of
the H.T. supply system, giving a rise of H.T. voltage. This is
applied to the phase-splitter and as the original rise of voltage
has caused a further rise, the action is regenerative. With an
odd number of states following the phase-splitter feed-back to
it is degenerative, but with an even number it is regenerative.
The two sides of a resistance-coupled amplifier, even apart
from the phase-splitter, operate in parallel as far as feed-back
effects are concerned. Consequently, the advantage of being
able to use little or no decoupling, which is found with trans-
former-coupled amplifiers^ is absent.
However, negative feed-back caused by the H.T. source
impedance does not necessarily affect the performance from the
point of view of the signal. In a single-sided R.C. amplifier,
it causes a drop in bass response, but in a push-pull R.C.
amplifier it does not, because under linear conditions the two
sides can operate in push-pull for the signal and in parallel for
feed-back voltages without interaction between the two.
Under non-linear conditions feed-back is liable to increase
amplitude distortion somewhat.
Positive feed-back is similarly harmless unless it is sufficient
to give rise to motor-boating. The valves are then driven
239
WIRELESS SERVICING MANUAL
Fig. i08 : In the paraphase form of phase-splitting circuit, there is a con-
siderable amount of self-balancing action
between current cut-off and grid current alternately and the
signal path is seriously affected.
In general, with Class “A'' and “small’* Class “ AB ”
output stages, it is satisfactory to omit decoupling in the
penultimate stage, but the stage before that must be heavily
decoupled. Thus, if Fig. io6 (a) precedes Fig. 102, the anode
circuits of Vi and V2 in the latter need not be decoupled, but
R4 and Ci in the former will have to be about 50,000 ohms
and 8 /uF. respectively. With “ heavy Class “ AB ” and
Class “ B ” stages the penultimate stage should be decoupled
also in order to prevent distortion arising through harmonics
of the signal being fed back.
240
PUSH-PULL AMPLIFIERS
Some phase-splitting circuits include a balancing adjustment.
Thus, in Figs. 104 and 105, Ri and R2 in the former and R3
and R4 in the latter may be a potentiometer with the output
taken from the slider. In Fig. 106 (a) Ri or R3 may be a variable
resistance, and in (b) Ri or R2 may be variable. Similarly, in
Fig. 107 Ri or R2 may be variable or they may be joined to
the ends of a potentiometer, the slider of which is connected
to H.T., while in Fig. 108 R6 and R7 are often a potentiometer.
The adjustment is easily carried out by feeding the amplifier
from an oscillator, preferably at about 400-1,000 c/s, and
adjusting the balance control for minimum output in a pair of
phones connected across a low resistance in the lead to the
centrc'-tap on the output transformer primary. As an alter-
native to this, adjust the input to give a convenient output in
the loudspeaker and then change the connections of the output
stage from (a) of Fig. 109 to (b) ; that is, transfer one of the
“ anode ” copnections on the output transformer to the other
“ anode ” terminal. The balancing control can then be adjusted
for minimum output. Ideally this would be zero but in
practice a trace of the signal may remain.
If the input is large, it will be found that there is a con-
siderable output at the balance point but the note is an octave
higher than the input. This is because the output is the
second harmonic of the input and, as the valves are not con-
nected in push-pull, it is not balanced out. When the balance
Fig. 109 : The change of connections shown at (b)
push-pull amplifier
241
is often useful when balancing a
WIRELESS SERVICING MANUAL
point has been determined, the connections to the transformer
can be replaced to normal.
A difficulty which occasionally arises with resistance-coupled
push-pull amplifiers is a lack of balance at very low frequencies,
usually well below audibility. None of the circuits described
is completely symmetrical. In Figs. 104, 105, 107 and 108
there is one more R.C. coupling in one side of the chain than
in the other, and in Fig. 106 there is the impedance of the
decoupling circuit on one side but not on the other.
By using suitable circuit values the lack of symmetry only
becomes apparent at very low frequencies and does not affect
the signal performance. It does sometimes, however, lead to
motor-boating through feed-back to an I.F. or R.F. valve.
Low-frequency voltages developed across the impedance of the
H.T. supply can produce motor-boating, which occurs only
when a signal is tuned-in, by modulating the carrier in an
R.F. or I.F. valve. Fortunately, it is a rare defec^.
The proper remedy is to decouple the H.T. supply to the
R.F. and I.F. valves, but this is rarely possible on account of
the heavy current which they draw. The best course, therefore,
is to reduce the gain at very low frequencies by reducing the
values of the coupling condensers and grid leaks as much as
possible without appreciably affecting the bass response.
When this is done the condensers and grid leaks in each stage
should be matched as closely as possible. The odd coupling
(C2, R3 in Fig. 104 and similar components in others) should
be made as large as possible to preserve symmetry.
Parasitic Oscillation
None of the circuits illustrated includes grid or anode
stopping resistances . 1 1 is customary to include such resistances
in push-pull amplifiers to prevent parasitic oscillation. They
are rarely needed anywhere but in the output stage, and not
always there.
With transformer coupling parasitic oscillation at a low
radio-frequency is theoretically possible with certain trans-
former characteristics. For it to occur it is necessary that the
leakage inductances of the transformers and the various stray
capacities to be correctly related, so that they form tuned grid
and tuned anode circuits with fairly close resonance frequencies.
If this occurs, the use of a grid stopper of some o*i megohm
for each output valve would be an effective cure, but the
author has never met a case.
242
PUSH-PULL AMPLIFIERS
Parasitic oscillation is usually at a very high radio-frequency
and can occur with either transformer or resistance coupling.
The valve and wiring capacities and inductances form resonant
circuits and turn the stage into an U.S.W. oscillator. The
best cure is to make the connections so short that at least one
of the circuits is resonant at a frequency above any at which
the valves will oscillate, and to make the frequencies of the
others very widely different.
This is not usually difficult, for it is generally easy to lay out
the apparatus so that all leads on the grid side of the output
stage are very short, and it often happens that the leads to the
output transformer are fairly long.
The second method of attack lies in damping the resonant
circuit by means of grid and anode stopping resistances. To
be effective they must be mounted directly on the valve holder
with very short leads. If care be taken in the wiring, as
described above and a 50-ohm stopper is connected in each
anode lead, trouble is very unlikely. If parasitic oscillation is
found in spite of these precautions, try io,ooo-ohm grid stoppers
as well.
With pentode output valves the screen circuits are equally or
more important, and 50-ohm stoppers should normally be con-
nected in the screen leads. The difficulties of parasitic oscilla-
tion are often greatly exaggerated, however, and if the layout
is good, most amplifiers will work well without stopping
resistances. The author usually includes them as a precaution
rather than of necessity.
The symptoms of parasitic oscillation are distortion and low
output. The anode currents will be markedly different from
the correct ones in most cases. With transformer coupling the
currents may be very high, and if the oscillation is allowed to
persist the valves will have only a short life. With resistance
coupling the currents are likely to be abnormally low. The
effect on valve life may still be bad, however, for there will be
grid current and some output valves are not built to withstand
appreciable grid current.
243
CHAPTER 24
MISCELLANEOUS DEFECTS AND ADJUSTMENTS
T he chief defects encountered in wireless receivers, and
the methods which are best adapted for tracing them, have
been described in detail in the preceding chapters. By
following the procedure laid down, the location of many faults
becomes almost an automatic process. More obscure troubles,
of course, may fail to submit to straightforward testing, but in
the majority of cases little difficulty will be found in devising
suitable tests and drawing the correct deductions from their
results. There is, however, a number of faults which, although
essentially simple in origin, prove puzzling to the tester, and
before concluding it is felt that a few remarks on defects of this
nature may prove helpful.
One of the simplest circuit arrangements is capable of pro-
viding symptoms which can be quite puzzling until one has
learnt to recognize them. Small straight sets are usually fitted
with reaction in order to obtain adequate sensitivity and selec-
tivity, and it sometimes happens that although the receiver
functions well on the medium waveband reaction fails to work
properly on the long waveband, or only functions over a portion
of the band. The disconcerting feature of this defect is that it
is quite possible to make the detector oscillate by increasing
the setting of the reaction control, but it fails to have any
appreciable effect on sensitivity and the characteristic whistle is
often absent when tuning through a station.
The trouble is only that the valve is oscillating at a frequency
determined by the constants of the anode circuit instead of the
grid circuit. It usually occurs with circuits similar to Fig. no,
when a single reaction coil does duty for both wavebands and
only the tuning coil in the grid circuit of the valve is switched.
On the long waveband, the reaction circuit sometimes takes
control and an increase in reaction has no effect on the signals
to which the grid circuit is tuned. When the reaction condenser
is advanced sufficiently, oscillation commences, usually with a
plop, but at a frequency determined by the reaction circuit.
A further advance in the setting of the control may cause the
oscillation to change over to the correct frequency of the grid
circuit, and the normal heterodyne whistles will then be
obtained.
The remedy for this state of affairs is usually very simple and
consists of nothing more than the insertion of a resistance in
244
MISCELLANEOUS DEFECTS AND ADJUSTMENTS
series with the reaction coil, so that the reaction circuit is
heavily damped and oscillation at any frequency determined by
it is prevented This resistance is shown at R in Fig. no and
its value should be as high as possible consistent with proper
reaction effects being obtained throughout the tuning range of
the receiver.
It will generally be found that a resistance of 500 ohms will
prove satisfactory, but in some cases it will have to be lower
in order to obtain proper reaction effects, and in others it must
be higher to prevent oscillation at the wrong frequency.
Apart from push-pull stages, balancing operations are not
often required nowadays. Occasionally, however, neutralised
R.F. amplifiers are met with in old receivers, and are likely to
require rebalancing, particularly if new valves are fitted. When
only one R.F. stage is used, it is possible to adjust the neutralis-
ing merely by setting the neutralising condenser to the point
at which stability is maintained throughout the tuning range.
A more refined procedure is necessary when the amplifier
includes more than a single stage, however, and is really advis-
able even when there is only one valve. The test oscillator
should be set to about 1,200 kc/s and its output connected to
the grid circuit of the stage preceding the one to be adjusted.
The neutralising condenser should be set at minimum and the
2 ^
WIRELESS SERVICING MANUAL
positive L.T, lead to the valve concerned disconnected (in the
case of indirectly heated valves, one of the heater leads). The
signal from the test oscillator should then be accurately tuned
in, and it will be found that sufficient energy will be passed by
the valve capacity for this to be readily done in spite of the
valve being inoperative.
The neutralising condenser should then be adjusted for
minimum output, and this setting should be clearly defined.
The filament or heater connections can then be replaced, and
the stage should be found to give definite amplification and be
quite stable.
The output of the oscillator can then be transferred to the
preceding stage and the next earlier valve in the chain neutralised
in the same way, the oscillator being finally connected to the
aerial and earth terminals while the first R.F. stage is adjusted.
The most widely used neutralised receivers were fitted with
only a single R.F. stage and there should be little difficulty
with any of these.
Two-R.F. sets were used fairly widely at one time, however,
and sets with as many as five R.F. stages have been produced
2 ^
MISCELLANEOUS DEFECTS AND ADJUSTMENTS
commercially. The difficulties of neutralisation are, of course,
much greater with these, and it may be remarked that in some
cases better neutralisation is secured when small resistances are
inserted in series with the neutralising condensers. It should
also be remembered that such circuits are often rather prone
to parasitic oscillation, particularly those types embodying
tapped tuned circuits.
Provision is made in most receivers for the connection of a
gramophone pick-up, but this is usually no more than a switch
which enables the pick-up terminals to be connected to the
A.F. volume control in place of the detector output. Apart
from a bad contact in the switch, therefore, there is nothing in
the average receiver which is likely to prove faulty on gramo-
phone but which does not show up also on radio.
Gramophone Pick-up
The pick-up itself, however, can develop faults which vary
with the type of pick-up. Apart from such obvious defects as
open-circuits in the windings and leads, which are readily found
with an ohmmeter, the usual faults are mechanical rather than
electrical. In most cases, the needle-holder and armature are
mounted in rubber supports; in the course of time the rubber
perishes, and the armature chatters against the pole pieces.
When the rubber is badly perished, the armature may be found
adhering to one of the pole pieces.
The remedy is, of course, to fit new rubber suspension, and
it is not difficult to do this in most cases if a little ingenuity is
exercised. It is necessary, however, to be careful to make the
suspension free enough. The suspension should be as free as
possible consistent with the needle's being able to support the
pick-up head on the record without bringing the armature into
contact with the pole pieces. Excessive stiffness means record
wear and, probably, resonance within the audible range.
The same principles of avoiding stiffness in the needle
mounting apply to other types of pick-up, such as the piezo-
electric, but here, of course, there is no question of an armature
and pole pieces. The needle-holder is linked to the crystal so
that its movement stresses the crystal and it produces an output
voltage by virtue of its piezo-electric property. Apart from
the rubber bushing of the needle-holder, the most likely defects
are in the mechanical linkage to the crystal and in the crystal
itself. The latter usually means a new crystal.
It should be pointed out that it is of no use testing a crystal
247
WIRELESS SERVICING MANUAL
pick-up for continuity, for the crystal is of infinite D.C. resist-
ance. One should, however, check the leads.
The tone arm itself should not be overlooked, for it can be
responsible for low-frequency resonances, chatter, and heavy
record wear. Special points to watch are joints and swivels.
Some tone anns have a rotatable head for easy needle changing.
This joint should be stiff to turn and when in the playing
position free from any trace of wobble.
All tone arms have joints of some kind which enable the head
to move both vertically and horizontally. Usually they are both
placed together at the pick-up mounting, but sometimes there
is only a rotating joint permitting horizontal movement at this
point, and the vertical movement is provided by a joint where
the pick-up head is attached to the tone arm.
Whatever the precise arrangement, the joints must be free
from any looseness, but at the same time they must be com-
pletely free. Any appreciable friction will cause record wear.
Some pick-ups are designed to have a steadily rising frequency
characteristic below about 500 c/s. This is done in order to
correct for the falling characteristic used in recording. Such a
pick-up characteristic is not an unmixed blessing, however.
Fig. M2 : Independent bsM and treble tone controls are obtainable with this
circuit. Care must be taken in positioning the treble accentuation»choke, for
it is very liable to pick-up hum
248
MISCELLANEOUS DEFECTS AND ADJUSTMENTS
for it is often achieved by the suitable placing of a bass resonance,
and this Usually means increased record wear. Often, too, the
increasing bass is accompanied by non-linearity.
With the better quality pick-ups, therefore, a flat charac-
teristic is aimed at, and it is then necessary to provide bass
compensation in the amplifier. This is quite easily done and a
first-stage A.F. amplifier like that shown in Fig. iii is very
satisfactory. The voltage amplification is about 3*8 times at
medium and high frequencies, and rises steadily below some
500 c/s.
A very good tone control stage can be built on these lines by
providing a switch to change the value of C i . Smaller capacities
than that shown will make the bass rise begin at higher fre-
quencies, and larger capacities will make it start at lower
frequencies. Replacing the capacity by an inductance will give
a flat bass, but rising treble characteristic; inductance values of
o *5-1.5 H. are suitable. Capacity and inductance in series
will make the response rise at both bass and treble.
Bass cut is easily arranged by providing a switch to reduce
C2, and treble-cut with another switch to shunt capacity across
the coupling. The switches can be linked to give two controls
providing independent bass and treble tone controls which have
a negligible effect upon the gain. The arrangement is shown
in Fig. 1 12 and five different degrees of bass response are
obtainable in any combination with five different degrees of
treble response. Twenty-five different frequency charac-
teristics are provided, therefore.
1
249
CHAPTER 25
NEGATIVE FEED-BACK
T he presence or absence of deliberate negative feed-back
in an A.F. amplifier makes very little difference in fault-
finding and troubles in the feed-back circuit itself are
confined almost entirely to such simple things as faulty resist-
ances. Its use is becoming common in the better broadcast
sets, and in a few cases it is bound up with tone control systems,
so that it may be as well to describe some of the principle
methods adopted.
There are two kinds of negative feed-back — current and
voltage. The former occurs when the voltage fed back to the
input is proportional to the output current and the latter when
it is proportional to the output voltage. Current feed-back
increases the apparent output resistance of a stage, while
voltage feed-back reduces it.
There is only one kind of current feed-back in common use
and this is the feed-back obtained from an unbypassed cathode-
bias resistance. The omission of the condenser reduces the
stage gain by the factor i/ -f where Rc is the
cathode resistance, /i and Ra are the amplification factor and
A.C. resistance of the valve, and Z is the anode circuit coupling
impedance. Where Ra is very large compared with Z, as in
pentode stages, the factor reduces approximately to i/(i + gRc)
where g is the mutual conductance.
The great advantage of the feed-back is that it improves the
linearity of a stage and so reduces amplitude distortion. Dis-
tortion is usually quite low in early A.F. stages, however, and
feed-back within such stages is rarely needed to reduce it. In
general, current feed-back is incidental to other requirements.
For instance, in one very common circuit to be described later
the feed-back is over two stages, and is applied to the cathode
of the first. This necessitates the cathode resistance of the
first stage being unbypassed, and so there is incidental current
feed-back from it.
If the gain of an A.F. amplifier is too great, but not so great
that a stage can be omitted, one or more bias resistances in the
early stages are sometimes left without bypass condensers . The
current feed-back reduces the gain, the linearity is improved
to some extent, and the condensers are saved.
250
NEGATIVE FEED-BACK
Heavy current feed-back occurs in the push-pull phase-
splitting circuit of Fig, io6, and makes the stage extremely
linear. The feed-back is so great that the voltage amplification
between the input and either of the two outputs is a little less
than unity.
In all these circuits the increase of apparent output resistance
resulting from the feed-back is unimportant. This is not the
case in the output stage. Here it is desirable to have a very
low output resistance in order to reduce distortion in the out-
put transformer. The use of current feed-back at this point,
although it may reduce valve distortion, tends to increase trans-
former distortion. The bias resistance of the output stage,
therefore, should always be shunted by a large capacity con-
denser, except when using push-pull or one of the circuits with
which grid de-coupling is permissible.
Negative voltage feed-back reduces the stage gain and
improves the linearity just as much as current feed-back, but it
Fig. 113: As explained in the text, it is usually better to feed-back from the
transformer secondary, as at (b), rather than from the primary
251
WIRELESS SERVICING MANUAL
reduces the apparent output resistance of a stage. It is, there-
fore, almost invariably employed in the output stage.
The voltage to be fed-back is taken from one of two places —
the anode of the output valve or the output transformer
secondary. The latter alternative is adopted wherever possible
and for two reasons ; no isolating condenser is necessary in
the feed-back path and the transformer winding resistances are
reduced by the feed-back. This is shown in Fig. 113 ; the
usual connections for feed-back from the valve anode are shown
at {a) and an isolating condenser C is needed to keep the H.T.
from the feed-back circuits. This is an additional component,
and is disadvantageous because it introduces some phase-shift
at low frequencies. The resistances ri and rz represent the
transformer primary and secondary winding resistances. From
the point of view of the transformer the effective resistance
across its primary is Ra -f ri without feed-back, and Ra/k -f- n
with feed-back.
With feed-back from the secondary as in (b)y however, the
effective resistance is (Ra -f- ri)/k. If the transformer resistance
is low, say under 200 ohms, and a moderate output resistance of
perhaps 1,500 ohms is required, the difference between the two
circuits is small. On the other hand, if the transformer resist-
ance is at all large, and especially if a low effective resistance is
needed, the difference can
be quite big. Thus,
suppose that the trans-
former resistance is
400 ohms and the
effective resistance across
the transformer primary
is required to be 700 ohms,
then with circuit (a) it is
necessary to reduce
by r feed-back the valve
resistance to 300 ohms
only. With circuit (6),
however, it is the valve
resistance plus 400 ohms
which must be reduced
to 700 ohms ; 400 ohms
is still negligible compared
with the valve resistance,
Fig. 114: With transformer coupling feed- SO that virtually the valve
back can be obtained over a aingfe nagc resistance mUSt be
by vaing thia orcuit
252
NEGATIVE FEED-BACK
brought down to 700
ohms instead of to 300
ohms. Less feed-back
will be needed, and the
gam will be greater. In
this case, for the same
effective resistance on
the transformer the gain
will be 7/3 times as
great with circuit (^) as
with circuit (a) .
When feed-back takes
place over the output
stage alone, it is often
necessary to feed-back
from the valve anode
because the voltage on
the transformer second-
ary is not as a rule
sufficient. The usual
circuit for a transformer-
coupled stage is shown
in Fig. 1 14 ; the voltage
amplification measured
between the input transformer secondary and the output trans-
former primary is A/(i + Afi) where A is the amplification
without feed-back and jS is the feed-back factor. In this case,
P ~ R2/(Ri + R2) at frequencies for which the reactance of C
is negligible. The output resistance Ro = Ra/(i + fiP).
As an example, take a pentode with g = 6mA/V. and an A.C.
resistance of 60,000 ohms. Suppose that the load is 10,000
ohms, and that an output resistance of 1,000 ohms is required.
Then = 59, and as /x = 360, P — 0*164. 'T'he amplification
without feed-back is 51*5 times, and with feed-back it is 5*45
times. Feed-back has reduced the output resistance to one-
sixtieth of its normal value and the amplification to a little less
than one-ninth.
For this amount of feed-back Ri is 5*1 times Ra, so that in
practice, one would make R2 about 30,000 ohms and Ri
some 150,000 ohms. A capacity C of 0*2-0 *5 /xF. would be
satisfactory.
A pentode of this type would normally need an input of
perhaps 10 volts peak ; with feed-back the voltage on the trans-
former secondary will be 51 volts peak. This must be supplied
253
Fig. 115 : This circuit is sometimes used and is
very simple, but has certain disadvantages, as
explained in the text
WIRELESS SERVICING MANUAL
without distortion by the preceding valve. With transformer
coupling, there is unlikely to be any difficulty in this, but the
fact well illustrates a point which must be carefully watched in
feed-back circuits. Feed-back reduces distortion in the circuits
over which it is applied, but those circuits require a correspond-
ingly larger input. The preceding circuits, therefore, must be
designed to supply this larger voltage without distortion. If
they are not, it is quite possible for the net effect of applying
feed-back to be to increase distortion.
When resistance coupling is used the circuit of Fig. 115 is
sometimes adopted and has the great merit of simplicity. The
preceding valve is shown because the amount of feed-back is
affected in some degree by its A.C. resistance. In general, this
penultimate valve must be a pentode, because the effect of
feed-back in this case is to make the apparent input resistance
of the output stage very low. The load into which Vi works
is quite small, therefore, and this is a bad condition for a triode
but a good one for a pentode. With any large amount of feed-
back serious difficulty will be found in obtaining sufficient un-
distorted output from Vi to drive the output stage. The circuit
is normally only useful when small amounts of feed-back are
wanted and it is not often used.
The most widely used of all feed-back circuits is that of
Fig. 1 16. It is a conventional two-stage amplifier with the by-
pass condenser omitted from the bias resistance in the first
stage, with one side of the output transformer secondary earthed,
and the other side connected to the cathode of Vi through R3.
If R3 is disconnected from the transformer and connected across
R2 the overall voltage amplification is A1A2, where Ai is the
amplification of the first stage measured between its grid and
anode and Az is the gain of the second stage similarly measured.
There is internal current feed-back in Vi from the cathode
resistance comprising R2 and R3 in parallel. The gain Ai is
measured or calculated taking this into account.
When R3 is connected to the transformer secondary as shown,
the voltage amplification becomes AiA2/(i -f AiAznp) where
P = R2/(R3 + R2) and n is the ratio of secondary/primary
turns on the transformer. If the normal output resistance is
Ra, the A.C. resistance of the output valve, with feed-back it
becomes Ro == Ra/(i 4- AiptnjS).
As an example, take the same pentode as before and assume
that Ai = 20 and that n = 1/40. Since Ra is 60,000 ohms, /i is
360 and Rois required to be 1,000 ohms, a simple sum shows
that /3 is 0*327. The amplification A2 is 51*5 ; therefore, the
254
NEGATIVE FEED-BACK
overall amplification with feed-back is 109 *4 times. With the
voltage feed-back removed it would be 1,030 times.
The values of R2 and R3 are chosen to fit two requirements
simultaneously ; they must be in the correct ratio to give the
required value of p and their value in parallel must equal the
bias resistance needed by Vi. If Rc is this required value of
bias resistance, then R3 = Rc/P and Rz = R^KilP — i).
Continuing the above example, suppose Rc is 2,000 ohms ; then
R3 must be 6,000 ohms and R2 is 3,000 ohms, in round figures.
The first valve Vi is often a pentode. When it is, its screen
should be decoupled to cathode and not to earth, otherwise it
will function as a pentode for the input voltage but a triode for
the fed-back voltage. The action is complex and not readily
amenable to calculation.
The frequency response of an amplifier such as that of Fig.
1 16 can readily be modified by including reactances in the feed-
back circuit. Thus if a condenser be connected in series with
R3 its reactance will increase at low frequencies and reduce the
feed-back ; consequently the amplification will increase. An in-
ductance in series with R3 will give a rising treble characteristic.
WIRELESS SERVICING MANUAL
A condenser in shunt with R3 will reduce its impedance at
high frequencies and increase feed-back, giving a falling treble
characteristic. Similarly an inductance in shunt with it will
give a falling bass response.
Artifices of this nature are sometimes adopted, but they should
not be used haphazardly, because the changes of gain which
they cause are reflected in changes in the output impedance of
the amplifier. There is always the possibility of some un-
desirable effects being caused thereby.
In all the circuits shown the output valve has been a pentode.
This is because negative feed-back is most needed with pentodes
owing to their high A.C. resistances. There is no reason why
it should not be used with a triode output valve if an exception-
ally low output resistance is required, but it is not usually
necessary.
Roughly speaking, negative feed-back with a pentode turns
it into the equivalent of a triode as far as amplification, output
resistance and linearity are concerned. It may be said, there-
fore, why not use a triode in the first place ? The answer is
that the pentode does give certain advantages ; in the first place,
when feed-back is over two stages, the penultimate stage has to
give a smaller voltage output and there is less risk of distortion
at this point, and in the second place the pentode has a higher
power efficiency. This means that it takes less power from the
H.T. supply for a given output power.
This comparison is between an indirectly heated pentode and
a directly heated triode. If the triode is indirectly heated also,
as it would have to be for an AC/DC set, the pentode scores
much more heavily. Indirectly heated triodes are not nearly as
good as directly heated types for output valves, and the triode
requires three to five times the grid bias of a pentode. In an
AC/ DC set the bias voltage is obtained at the expense of the
anode voltage, for the H.T. voltage is fixed. A triode would
therefore give a much smaller output than a pentode.
To sum up, in AC/DC sets a pentode output valve with
negative feed-back is greatly superior to a triode. In A.C. sets,
where there is little or no limitation on the H.T. supply there
is very little to choose. The pentode with negative feed-back
is slightly more efficient and has certain slight theoretical
advantages, but the triode circuit is unquestionably simpler.
With big A.C.-operated qutput stages, however, the power
efficiency becomes important and then the pentode with
negative feed-back is usually to be preferred. Such stages are
usually push-pull, however, and usually Class “ AB.”
CHAPTER 26
TELEVISION RECEIVERS
W HEN television ceased at the start of the war the num-
ber of television receivers in use was small compared
with the number of broadcast sets and any defects
which developed were usually attended to by their makers.
Nevertheless, the number of sets was increasing and when tele-
vision starts again in the future the professional service-man
will be called upon to tackle a new set of problems ; the
amateur also, in his experimental work, is liable to meet with
defects which must be removed. It is felt, therefore, that
some notes on the particular problems of television may be
helpful, and the following remarks are based upon the author's
experience with a wide variety of equipment .
Defects which may develop in equipment which has been
working satisfactorily will be considered first. Their location
and cure usually involve no great difficulty, for television equip-
ment obeys the same general laws governing all wireless
apparatus. Since the defects manifest themselves visually
rather than aurally, however, diagnosis is rather more trouble-
some at first. The sound receiver needs no attention here, for
any faults in it can be treated in the same way as in any other
sound receiver. The vision receiver, too, is most easily attended
to by connecting a pair of phones to its output, for then signals
are audible and the results can be judged in the familar way.
In general, the possible defects in the vision receiver are the
same as those which may develop in any sound receiver and are
treated in the same way. There are, however, certain faults
which give rise to no audible sound but which have a marked
effect on the picture. A fault which causes the appearance of a
series of faint diagonal lines across the picture, for instance,
gives rise to no audible effect in a pair of phones because the
frequency of the output current responsible is well above
audibility. This fault may occur in a superheterodyne and is
normally due to harmonics of the intermediate frequency being
fed back to the input circuits. The effect is, in fact, exactly
the same as I.F. harmonic feed-back, which is dealt with in
Chapter 14 for the case of broadcast receivers.
In the case of a new receiver which is being tried out, such
feed-back is nearly always a sure indication that the screening
is insufficient or that the filter after the detector is inadequate.
The precise intermediate frequency used has also an important
257
WIRELESS SERVICING MANUAL
Fig. 117 ; Resonance curves for different waysof tuning ; (a) isfor all circuits
tuned to the same frequency, fb) is correct with the circuits staggered, and
(c) shows Incorrect adjustment
bearing on the magnitude of the effects. With a receiver which
has been operating correctly, however, it may usually be taken
that the appearance of this fault is a sign that a defect has
developed in one of the components in the detector filter or,
more likely, that a bad contact has made its appearance in some
of the screening. Incidentally, it should be noted that parasitic
oscillation in almost any valve is likely to give very similar
symptoms.
Interference from Sound Transmitter
These diagonal lines should not be confused with the hori-
zontal lines which may be produced by interference from the
sound transmitter. This interference is usually readily dis-
tinguishable because it varies in intensity, and in the number
and width of the lines, in sympathy with the sound modulation.
The trouble is due to a lack of selectivity in the vision receiver
or to cross-modulation in an early valve. The latter is generally
caused by operating the apparatus with too large an input, but
mis-alignment of the early tuned circuits towards the sound
transmitter can also cause it. The defect is more likely to occur
in apparatus in which the early stages are common to both
sound and vision channels than in equipment having entirely
separate sound and vision receivers.
The precise tuning of the R.F. and I.F. circuits as well as the
setting of the oscillator condenser has a marked effect upon
picture quality. With a straight set the circuits are usually very
heavily damped by the input resistance of the valves, for at
45 Mc/s the electron transit time is important ; as a consequence
tuning is very flat and has little effect upon the picture quality,
unless a certain amount of regeneration is present. There is
always some regeneration due to stray couplings, consequently
258
TELEVISION RECEIVERS
one or more of the circuits may have a considerable effect upon
the definition.
If the circuits are too sharp and are tuned to the same fre-
quency, the definition will be spoilt and the picture will appear
fuzzy. This effect should not be confused with the fuzziness
caused by poor focusing of the light spot on the tube. In the
former case the lines will be quite clear and sharp, but in the
latter they too will be fuzzy or even indistinguishable.
When moderately sharp circuits are used, the usual practice
is to mistune pairs of circuits slightly in opposite directions to
broaden the resonance curve. When this is done, the precise
settings of the trimmers are usually quite critical. It may
happen, for instance, that with all circuits in tune the resonance
curve has the form of Fig. 1 17 (a) ; by judicious mistuning of the
circuits it can be brought to the more desirable shape shown
at (b). If the mistuning is improperly carried out, however, the
curve may take the form of (r), with rather a curious effect on
the picture.
What is known as transient distortion usually occurs. A
sudden change in current such as that shown in Fig. 118 (a) is
not reproduced properly but is distorted to the form shown at (b).
The visible effect is that immediately to the right of a white
object there is a narrow black band followed by a white band,
and then another black band and so on. The number of bands,
their width and intensity, depend upon the characteristics of the
receiver and upon the precise shape of the resonance curve.
More than two bands are not common.
In a superheterodyne, incorrect adjustment of the R.F. or
I.F. circuits will cause the same effects. The precise setting of
the oscillator tuning condenser also has a considerable e&ct
upon the picture quality, especially if the receiver is of the
single-sideband type. When the control is set so that the
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WIRELESS SERVICING MANUAL
carrier falls in the centre of the I.F. resonance curve the picture
detail is below normal ; as the control is rotated so that the
carrier comes to one edge of the pass-band the definition im-
proves and it is usually easy to find an optimum setting for picture
quality. This setting usually gives about half the output
obtained with the carrier on the peak of the resonance curve.
If the oscillator control is set so that the carrier falls well
outside the pass-band, not only will there be a big loss of signal
strength, but a peculiar form of distortion will appear. This
is known as “ plastic *’ ; most of the light and shade of the
picture disappears and objects acquire strong outlines, having
the effect of being in relief. It is caused by an almost complete
lack of the lower modulation frequencies coupled with strong
reproduction of the higher frequencies.
In general, when plastic or multiple images occur in a receiver
which has previously been free from these effects, it can be
taken that the tuned circuits require re-alignment. It should
be pointed out, however, that these effects may be caused by
faults present in the signal path outside the receiver. Reflections
can occur in an incorrectly terminated aerial feeder which cause
similar distortion of the picture.
The trouble may also arise in the path of the signal between
the transmitting and receiving aerials. The received signal
may have travelled by several different paths ; one may be
direct, the others will be indirect and be present owing to
reflection from buildings or other objects. The signals coming
by the indirect paths take longer to reach the receiving aerial
than those coming by a direct path and so give rise to multiple
images.
The remedy lies in so arranging the aerial that it is responsive
to the signal from one path only. A highly directive aerial is
obviously required and it must be oriented experimentally to
the optimum direction. In the author’s experience, however,
these external causes of multiple images do not cause much
trouble at normal distances from the transmitter. It is, never-
theless, well to bear their possibility in mind when distortion is
evident and it cannot be remedied by the adjustment of the
receiver.
Unfamiliar Defects
The defects which, because of their unfamiliarity, are mostly
likely to be troublesome, however, lie in the time-base and
cathode-ray tube circuits, and here a distinction must be
made between electric and electromagnetic deflection and
260
TELEVISION RECEIVERS
focusing, for the associated equipment is different in the two
cases. With electric deflection four deflecting plates are built
into the tube and deflection of the electron beam is obtained
by applying voltages of saw-tooth wave-form to these plates.
The horizontal plates have a voltage of 10,125 c/s applied, and
the vertical deflection plates one of 50 c/s. The amplitude of
the deflecting voltages is usually about 1,000 volts peak-to-peak.
If the whole illuminated area of the screen is to be evenly
focused and if there is to be no interaction between the two sets
of deflecting plates, it is necessary for the mean potential of
each pair of plates to be constant and independent of the
amplitude of the deflecting voltage. This necessitates the
deflecting voltages being applied in push-pull to the plates. The
conventional time-base for electric deflection thus consists
of a gas-filled triode saw-tooth oscillator and a two-valve triode
paraphase-type amplifier. An H.T. supply of about 1,000 volts
is usually adopted.
Special Requirements
The circuits employed largely follow ordinary amplifier
practice, but the frequency response and phase-shift require-
ments are different from those needed in a sound amplifier.
The frame time-base amplifier must give negligible phase-shift
over a frequency-band of about 50-500 c/s at least, while the
line time-base amplifier must be good over the range of approxi-
mately 10,000-100,000 c/s. In practice the only difficulty en-
countered lies in obtaining an adequate response in the line
amplifier, for stray capacities prove troublesome. The effect
of an inadequate high frequency response is to make the fly-
back time too slow’ and in the picture this can be seen as a cutting
off of the left-hand side of the picture. In extreme cases, the
left-hand side may be folded back on itself. The cause of a
slow fly-back time does not necessarily lie in the time-base,
but this is the first place to look and every endeavour should be
made to keep the capacity of the connecting leads to the line
deflecting plates at a minimum. The grid circuit of the para-
phase valve is also important from this point of view. Trouble
with a slow fly-back is rarely encountered in the frame deflecting
circuit.
Other causes of a slow fly-back are a defective gas-triode, or
more likely, serious deformation of the shape of the line syn-
chronising pulse. This is normally due to excessive stray
capacity in the circuits of the sync separator.
The output of the time-bases is coupled to the deflecting
261
LINE
DEFLECTION
WIRELESS SERVICinq MANUAL
262
electric focusing and deflection.
TELEVISION RECEIVERS
plates by condensers and resistances exactly in the manner of
an R.C. intervalve coupling. When the positive terminal of
the high-voltage tube supply is earthed, the potential across these
condensers is of the order of 600 volts only. When the negative
terminal is earthed, however, it is some 3,000-4,000 volts.
Whatever the connections, a leak in one of these condensers
will cause the position of the picture on the end of the tube to
alter, for one of the plates will receive a bias voltage with respect
to the one opposite it. A slight leak can be corrected by adjust-
ing the appropriate shift control which is provided for centring
the picture. Such controls operate by biasing the plates and if
a leak develops its correction by this means will alter the mean
potential of the deflecting plates. This will affect the focusing
and it may not be possible to refocus properly by adjusting the
second anode voltage. Thq correct thing to be done is to replace
the defective condenser.
Intermittent Condenser Breakdown
Instead of, or as well as, developing a leak, a condenser may
show an intermittent breakdown. When this occurs the picture
will fly off the end of the tube and then come back again slowly.
A frame deflecting circuit condenser will cause a vertical
movement of the picture and a line circuit condenser a hori-
zontal movement. The speed of return of the picture depends
upon the circuit values, but in the case of the frame deflection
may be about i or 2 seconds.
The tube is usually operated at about 4,000 volts and a
typical network is shown in Fig. 119. Here a half-wave
rectifier valve is used with a reservoir condenser Ci and a
resistance-capacity smoothing circuit Ri C2. The con-
denser Ci must be rated for working at the peak voltage of the
transformer, that is, \/2 times the R.M.S. transformer voltage,
and the rating of C2 should not be far short of this figure. A
typical transformer may be wound for 3,000 volts, and the
rating of Ci must then be not less than 4,250 volts. During the
non-conducting half-cycles the voltage across the valve may
reach twice this figure, or 9,000 volts. This is the peak inverse
voltage and is 2 times the R.M.S. transformer voltage.
It is very important to notice that this voltage appears between
the rectifier filament winding and one end of the high-voltage
winding and a very high standard of insulation is necessary if a
breakdown is to be avoided. Other arrangements of the
rectifier are possible, but the peak inverse voltage always
appears between some parts of the transformer with a valve
263
WIRELESS SERVICING MANUAL
rectifier and a negative earth to the high voltage supply. With
a positive earth it is possible to avoid this voltage’s appearing
in the transformer, and with either a positive or negative earth
it can be avoided with a metal rectifier.
After the smoothing, a potentiometer is connected across
the H.T. supply and the various electrodes of the tube are fed
from tappings on it. Measuring voltages with respect to
cathode, the grid is made about 30-200 volts negative and the
precise bias is adjustable by R8. This is the brilliancy control.
The first anode is about 250-400 volts positive and the second
anode about i ,000 volts positive ; its potential is adjustable by
R5 and is varied for focusing the light spot on the screen. The
third anode potential is fixed and is some 3,000-4,000 volts
positive.
The condensers and resistances C4, C5, C6, C7, Rio, Rii,
Ri 2, R13 are the coupling components from the time-bases to
the deflecting plates. Each of the leaks is returned to a poten-
tiometer (R14, R15, R16, R17) across the resistances R2 and R3
to the centre-point of which the third anode is returned. As a
result, the potential of any deflector plate can be varied positively
or negatively with respect to the third anode by perhaps 100-200
volts.
It is not usual to find four potentiometers connected in this
way. Normally two only are used and the two other leaks are
returned directly to the third anode. The two potentiometers
then form the horizontal and vertical shift controls and enable
the picture to be centred.
It has been found, however, that there is an optimum mean
potential for each pair of plates and that this may be positive or
negative with respect to the third anode. Much depends on
the particular tube used, of course, but it is often possible con-
siderably to improve the focusing, and hence the definition, by
using the four controls shown in Fig. 119. In this diagram
R16 and R17 are horizontal shift controls and R14 and R15 are
the vertical shift controls. By altering R14, the picture will
move ; an adjustment to R15 will bring it back again to the
centre. The position of the picture will then be the same as
before, but the mean potential of the plates will be different and
this will affect the focusing.
When adjusting a system of this type, first of all centre the
raster on the tube and then focus as well as possible with R5.
The next step is to determine whether better focusing can be
secured by an alteration in the mean potential of the plates and
the adjustments should preferably be carried out on a picture
264
TELEVISION RECEIVERS
rather than with a blank raster. The reason for this is tliat with
incorrect adjustments it is possible for the light spot to be
drawn out into a line, and if this is horizontal, good focusing can
be obtained as judged by the line width, but picture definition
will be very poor.
Changing the Mean Plate Potential
To change the mean potential of the plates, shift the picture
vertically two inches or so by R 14, centre t again by R15, and
then refocus by R5 . If the definition has improved or the focus
is more evenly maintained over the raster, try shifting the mean
potential more in the same direction. On the other hand, if the
results are worse it is clear that the mean potential is being
changed in the wrong direction. When the best results have
been obtained with R14 and R15, the horizontal controls should
be similarly treated. Do not forget to readjust R5 every time
one of the shift controls is altered.
This process is undoubtedly tedious, but it is well worth
while, for it is often possible to secure a really big improvement
in definition with careful adjustment.
When electromagnetic deflection and focusing are used the
circuits are quite different. The tube usually contains only
cathode, grid and anode, focusing and deflection being obtained
by magnetic fields produced with the aid of external coils
mounted around the neck of the tube. With correctly designed
coils it is easier to obtain even focusing over the raster than with
electric arrangements, but it is usually necessary for the
focusing coil to be carefully centred so that its axis is coincident
with the axis of the electrode assembly in the tube.
Focusing is accomplished by varying the current through the
coil by a variable resistance. As the resistance of copper wire
increases with temperature the current through the coil, and
hence the focusing, tends to alter as the apparatus warms up.
Normally this is not serious and if the adjustment is made with
the gear at normal temperature, the correct focus is reached a
few minutes after switching on.
Time-Base Output Requirement
In general, a time-base for magnetic deflection contains a
saw-tooth oscillator and a pentode amplifier. The output
required is in the form of a saw-tooth current wave ; the valves
are consequently operated at about 250 volts only but with a
large standing anode current. In the case of the frame deflect-
ing circuit, little difficulty arises, and an output valve taking
265
WIRELESS SERVICING MANUAL
30 mA or so can provide an adequate output to the deflecting
coil, which is usually choke-capacity fed.
The line circuit is much more difficult and it is the general
practice to use comparatively few turns on the deflecting coils
and to feed them through a step-down transformer. Even then
the output valve must take some 70 mA anode current if full
deflection is to be secured.
When the input saw-tooth voltage to the line output valve
has a rapid fly-back, the effect in the anode circuit is the same
as if this circuit were interrupted. Consequently a high back
E.M.F. is generated in the coils. Its value depends on the
current and the speed of fly-back, but it can easily reach several
thousand volts. This is one reason for using relatively few
turns on the line deflector coils ; a moderate number of turns
reduces the back E.M.F. on the coil and there is more space
available for adequate insulation.
The back E.M.F. across the coil, however, is stepped up by
the output transformer and a high peak voltage is applied to the
anode of the output valve on the fly-back. A resistance-con-
denser combination is often connected across the primary of this
transformer in order to limit the voltage, but even so it is
necessary to watch the insulation of this circuit carefully.
In particular, the valveholder should be carefully chosen.
The author met a case in which an arc developed between the
anode and two other sockets ; at first it was evident only when
the time-base was operating, but at length it became continuous
with no input to the valve. Examination of the holder showed
that a carbon track had appeared between the sockets ; it was,
of course, deposited by the initial high-voltage arcing. If a
fault of this nature occurs, the only thing to do is to replace the
holder and to see that the new one is of a type giving a maximum
clearance between the metal parts, especially the parts contacting
with the surface of the insulating material.
The general arrangements of the tube circuits is shown in
Fig. 120 and is obviously much simpler than the corresponding
equipment for electric deflection. No smoothing is usually
needed in the high voltage supply, a reservoir condenser Ci
being sufficient ; Rz is the bias control for brilliancy. All the
rest of the apparatus, including receiver and time-bases, can be
run off a common supply of the order of 250-300 volts only.
Mains hum in television equipment varies in its visible effects
according to its point of entry. With 50 c/s mains its effects
are stationary because the frame scanning is at the same fre-
quency. With an unsynchronised raster, however, the visible
266
TELEVISION RECEIVERS
Fig. 120 : In this diagram is given a common arrangement for a tube with
electromagnetic focusing and deflection. R2 is the brilliance control and R.5
the focus. Shift controls are often omitted
effects of hum will be moving vertically. Excessive ripple on
the H.T. supply of the line time-base causes the vertical edges
of the picture to be wavy instead of straight. The effect is quite
distinctive and its presence is proof that there is a ripple on the
line scanning voltage or current ; the precise cause of the ripple
must be found by searching in the appropriate circuits in the
usual way. Hum in the frame time-base causes less distinctive
visible effects ; usually, however, it upsets the interlacing. Its
presence can often be detected by the possibility of locking the
frame time-base at mains frequency without any sync pulse
from the signal.
Hum in the tube voltage supplies or in the receiver does not
affect the shape of the raster, but gives rise to a dark band
across the picture. With severe hum the band may be black ;
with slight hum it will be no more than a slight darkening of a
portion of the picture. Hum in the receiver may also affect
the frame synchronising, since it will get through the sync
267
WIRELESS SERVICING MANUAL
separator in some degree. Hum in the high-voltage supply,
however, will rarely have any effect on the synchronising.
Synchronising troubles are not ones which are very likely to
develop. When trying out an experimental receiver, of course,
all sorts of faults may be found, but in equipment which has been
operating correctly little is likely to go wrong. A complete
failure of synchronising may occur, through the failure of the
valve used for sync separation ; this is easy to diagnose, but
when checking its operating voltages do not forget that it is
normal with some types of amplitude filter to operate a screen-
grid valve with a normal screen voltage but some 6-io volts
only for the anode supply.
Those accustomed to high voltages will naturally be wary
of handling the equipment when it is switched on and will
remember that condensers can hold a charge for quite a long
time. In the high-voltage circuits it is customary for the con-
densers to have high resistances across them, so that they will
discharge if given time. Their discharge is quite slow, however,
judged by broadcast receiver standards, and a nasty shock can
be obtained if one puts one’s fingers across the terminals even a
minute or so after switching off. Always allow several minutes’
interval before poking about in the high-voltage circuits,
High-Voltage Faults
The high-voltage supply can be responsible for some very
elusive faults. The author had a case of a receiver which pro-
duced good results for some months and then started to give
trouble occasionally. The trouble took the form of the appear-
ance of bright spots on the picture, rather like car ignition inter-
ference, accompanied by poor synchronising. The trouble was
intermittent, but gradually got worse, and the interior of the
cabinet smelt strongly of ozone. This gave the main clue, for
ozone is produced by sparking. No sign of trouble appeared
in the high voltage unit, but it was at length noticed that faint
sounds of sparking could be heard at the base of the tube.
Upon removing the base from the tube and switching on, the
cause of the trouble was clear, for on looking into the base in
the dark, bright blue spots were evident on the corners of the
contacts. The trouble was caused largely by surface leakage
across the bakelite and corrosion of the contacts, and these in
turn were due to moisture in the atmosphere being deposited
on the bakelite and metal and carrying with it various corrosive
elements. The discharge, once started, made matters worse.
The remedy, of course, is to clean the tube base thoroughly
266
TELEVISION RECEIVERS
and to polish the contacts so that there are no sharp edges or
rough parts on the metal. Any roughness greatly increases the
likelihood of a discharge.
Corona may occur elsewhere in the high-voltage circuits and
may persist for a long time without apparently doing any harm.
Eventually, however, there is likely to be a breakdown, so that
it is as well to trace the source of ozone whenever it is smelt.
The most likely points of trouble are the rectifier, mains
transformer, . and the connections to these components ; the
trouble can arise in any part of the high-voltage circuits, how-
ever, where the insulation is poor or air-spacing is inadequate,
especially if the metal parts are not polished and free from sharp
edges.
Hunts Capacitance and Resistance Analyser
269
CHAPTER 27
THE USE OF CATHODE-RAY TEST GEAR
L ittle stress has been laid on the use of the cathode-ray
tube in receiver testing, for it is felt that even to-day it is
not generally available. While it costs much less than it
did some years ago, a good oscilloscope is still fairly expensive
and it is supplementary to other apparatus. By using it one
cannot escape the need for the ordinary test gear which has been
frequently referred to in earlier chapters.
There is no doubt that the cathode-ray tube is essential to
research and receiver design and it is almost essential to tele-
vision servicing, but 'it is far less important for broadcast
receiver servicing. This does not mean, however, that it
cannot be of very great assistance in this connection. With
suitable auxiliary apparatus it can be very valuable, and in cases
where large numbers of receivers are dealt with its cost can
readily be justified.
The main use of the oscilloscope in receiver servicing is with
a “ wobbly oscillator ” for the visual alignment of the various
tuned circuits. When l.F. transformers of the “ over-coupled’*
variety are used this equipment provides the only simple
method of adjusting them.
The usual commercial oscilloscope consists of a cathode-ray
tube, amplifier, linear time-base and power supply. The out-
put of the time-base is applied to the horizontal, or X, plates of
the tube and it moves the spot steadily across the screen, usually
from left to right. The fly-back is quite rapid and successive
traversals of the screen coincide with one another.
The number of traversals a second is variable by the operator
and the range provided is often from 2 or 3 a second to 5 ,000 or
so. Above about ten a second the eye cannot follow the spot
and there is the appearance of a line on the screen. Flicker
disappears above some 30-40 traversals a second.
The waveform to be examined is applied to the vertical or Y
deflecting plates. If its amplitude is insufficient for direct
observation, as is often the case, an amplifier is used and is
usually included within the case of the oscilloscope.
As it stands, the oscilloscope is useless for circuit alignment,
for it requires certain associated equipment, but is quite suited
for waveform examination. The waveform in all parts of an
A.F. amplifier, for instance, can readily be examined.
The amplifier should be provided with an input of good sine
waveform derived from some form of A.F. oscillator, and the
270
THE USE OF CATHODE-RAY TEST GEAR
input of the oscilloscope clipped in turn to each part of the
amplifier which it is desired to investigate. The amplifier input
should be kept constant while doing this and the desired size of
trace on the tube obtained by suitably varying the gain of the
oscilloscope amplifier.
If the trace departs noticeably from a sine waveform when it
is known that the amplifier input is good it is a sign of serious
amplitude distortion. The oscilloscope used in this way is not
a very sensitive detector of amplitude distortion, and if the input
is increased until the overload point of the amplifier is reached it
will usually be found that audible and visible distortion occur
together.
Where the oscilloscope scores over all other testing devices
is the readiness with which certain types of parasitic oscillation
can be detected. It sometimes happens that an amplifier may
be quite stable with its normal voltages, but go into oscillation at
a high frequency when the bias on one of the valves is reduced
nearly to zero.
Now in operation at full output the valve is operated at this
point every positive half-cycle of the input signal, w'ith the
result that parasitic oscillation occurs during part of every
positive half-cycle on the valve responsible. Normally this,
fault is very difficult to detect, for often the only symptom is
that distortion sets in at an abnormally low output.
With no input or only a small input, the amplifier will appear
normal in all respects and even with full input the change of
anode current is no more than would occur if the distortion
were due to other causes. When an oscilloscope is used, how-
ever, the trouble is immediately evident, for instead of the trace
of Fig. 1 21 («) a picture such as that of (b) is obtained.
Here the valve starts to oscillate at radio-frequency when its
271
WIRELESS SERVICING MANUAL
grid potential swings to zero on the A.F. signal and continues to
do so until the grid voltage falls to about half-way between zero
and normal bias. Oscillation then ceases and the remainder of
the A.F. cycle is dealt with properly.
When used with sine wave input voltages little difficulty is
usually experienced in synchronisiijg the time-base to give a
steady image. In television work, however, difficulty is often
experienced. The oscilloscope is used very largely for examin-
ing the waveform of the sync pulses in the various stages and
the effectiveness of the amplitude filter. It is found in practice
that sometimes there is no trouble with oscilloscope synchronis-
ing while at others there is.
The reason for this lies in the polarity of the signal applied to
the oscilloscope. Most types need a voltage change in a positive
direction to trigger the time-base,, and when the oscilloscope is
connected to a point in a television receiver at which the pulses
are in this sense no difficulty is experienced. It is when the
pulses are in the opposite sense that it becomes very trouble-
some to secure a stable trace on the tube.
It will be found, too, that the amplifier included in many
oscilloscopes is not good enough for television and that it very
^seriously distorts line sync pulses, so that no useful results can
be secured with it. The best course is to cut out the internal
amplifier and to build an external one with a response up to at
least 200,000 c/s and preferably higher.
Such an amplifier should be built on the lines of a television
V.F. stage and television type R.F. pentodes can well be used. It
is advisable also to include a phase-reversing stage so that the syn-
chronising difficulty referred to above can readily be overcome.
A suggested arrangement is shown in Fig. 122. Here Vi is
a phase-reversing stage, for the switch Si permits the output to
be taken from either its anode or its cathode. Ri and R2
should be equal and can well be 10,000 ohms when Vi is a triode
of about 4,000 ohms A.C. resistance. The gain is slightly
below unity and when the output is taken from the cathode both
input and output are in the same phase. Whatever the phase
of the input, positive sync pulses can be applied to the oscillo-
scope time-base.
The valve V2 is the amplifier and a gain of 20 -30 times should
be possible with a television-type R.F. pentode. Circuit values
should be worked out in accordance with the usual television prac-
ticeand depend largely upon the input capacity of the oscilloscope .
Every effort should be made to keep this capacity at a
niinimum, since for a given response R3 is inversely proportional
m
THE USE OF CATHODE-RAY TEST GEAR
Fig. 122 : An oscilloscope amplifier suitable for television work is shown here ;
VI is a phase-reversing stage and V2 is the amplifier proper
to the total circuit capacity, and the gain and undistorted output
of the amplifier are proportional to R3. Too high a capacity,
therefore, means that R3 must be too low for reasonable stage
gain and output.
It is normally only for television servicing that such a special
oscilloscope amplifier is necessary, and in the great majority of
cases encountered with sound apparatus the ordinary amplifier
is quite satisfactory. There is, however, one point of difficulty
which is quite likely to arise when dealing with transformer-
coupled A.F. amplifiers.
Most saw-tooth oscillators pass heavy grid current on the fly-
back and this sets up a sharp voltage pulse at the sync input
terminals. If these terminals are connected, in parallel with the
deflector plates or amplifier input, to a transformer or choke in
the amplifier under test this circuit will be kicked into oscillation
at its natural frequency and the trace on the tube will be seriously
distorted .
The effect will be absent only if the resonant circuit formed
by the inductance of the transformer or choke and the circuit
capacity is damped beyond a certain value. It will nearly always
be found if the oscilloscope is connected to the secondary of an
intervalve transformer unless that transformer has a damping
resistance across its secondary.
The effect can easily be avoided, by feeding the sync terminals
273
WIRELESS SERVICING MANUAL
in pmrallel with the deflector plates from the output of the
oscilloscope amplifier, so that this amplifier acts as a buffer
stage between the circuit under test and the time-base. No
trouble will then be encountered.
What is one of the most important uses of the oscilloscope in
servicing must now be dealt with. This is its application to the
visual alignment of tuned circuits. In addition to the oscillo-
scope a “wobbly oscillator ” or “ wobbulator as it is frequently
termed, is needed. Actually, this is an R.F. oscillator frequency-
modulated by the saw-tooth wave of the oscilloscope time-base.
The usual practice is to have an R.F. oscillator of any con-
ventional type operating at a convenient fixed frequency — say
3 Mc/s. Across the tuned circuit of this oscillator is connected
a frequency-control valve which functions in the same way as
the control valve in an automatic frequency control system.
It is a valve connected so that its output impedance is chiefly a
reactance, the value of which is a function of grid bias over
an adequate range. By changing the bras on the valve its output
reactance changes and hence the resonant frequency of the
oscillator circuit with which it is in parallel. The oscillator
frequency can thus be controlled by altering the grid voltage of
the control valve.
This valve has applied to its grid a saw-tooth voltage wave
taken from the oscilloscope time-base, so that the net result is
that the oscillator frequency varies with the time-base voltage.
The variation is usually about ± 30 kc/s from the mean value.
When the C.R. tube spot is on the left-hand side of the screen
the oscillator frequency may be 2*97 Mc/s and it will rise steadily
as the spot moves across the screen. In the middle it will be
3 Mc/s and when it has reached the right-hand side the fre-
quency will have become 3 *03 Mc/s. The fly-back then occurs;
the spot rapidly goes back to the left-hand side and the frequency
rapidly returns to 2*97 Mc/s. Distance horizontally becomes a
measure of frequency.
If the output of this frequency-modulated oscillator is applied
to the input of an amplifier tuned to the mean frequency, in
this case 3 Mc/s, the D.C. detector output will vary in accord-
ance with the gain of the amplifier at the frequency applied at
any instant. It will, in fact, vary according to the overall
resonance curve of the amplifier.
This direct current sets up a similarly varying voltage across
the resistances of the detector circuit and if this is applied to the
vertical deflector plates of the oscilloscope the position of the
spot in the vertical direction will correspond to the gain of the
274
THE USE OF CATHODE-^RAY TEST GEAR
amplifier at the frequency applied at that moment. The spot
consequently traces out the resonance-curve of the amplifier.
If the process is repeated sufficiently rapidly, the resonance-
curve becomes in appearance a solid line because of the reten-
tivity of the eye, and the changes of direct voltage in the detector
output can be passed through a resistance-capacity coupling.
This is important, since it permits an amplifier to be used
between the detector and the deflector plates, and so a reason-
ably large image can be secured.
As described so far, the oscillator has a fixed mean fre-
quency and it cannot be varied directly without affecting the
amount of the frequency modulation. To obtain an output at
any desired mean frequency the superheterodyne principle of
frequency changing is adopted.
Another unmodulated oscillator covering a wide range of
frequencies is used and the outputs of the two oscillators are
applied to a mixing stage — the difference frequency in the output
being applied to the apparatus under test. Thus, if an output
of 465 kc/s is wanted and the frequency-modulated oscillator has
a mean output at 3 Mc/s, the additional oscillator is set at
3 ± 0*465 = 2*535 or 3*465 Mc/s.
As the “ wobbly oscillator and the horizontal sweep are
both controlled from the time-base synchronism is inherent.
It is, however, usually advisable to lock the time-base to the
mains frequency, or a sub-multiple of it, to prevent any hum
from causing a wobble on the image. In spite of the inherent
synchronism the time-base frequency is quite important, and
it must be chosen correctly if a distorted and misleading trace
is not to be secured.
If the frequency is too high in relation to the selectivity of the
amplifier serious distortion of the trace will result, the distortion
occurring just beyond the peak of the resonance curve. The
better the circuits used in the amplifier, the lowermustbe the time-
base frequency and in no circumstances should it exceed 50 c/s.
The general practice is to use 25 c/s, but in some cases 12^ c/s
is better. Flicker of the image is then very noticeable, but is
not a serious drawback.
If the frequency is too low in relation to the characteristics
of the circuits between the detector and deflector plates, the
resonance curve will again be distorted. This will take the
form shown in Fig. 123 where the solid line curve indicates the
correct trace of the true resonance curve and the dotted line
shows the distorted trace.
This trouble is caused by an inadequate low-frequency
275
WIRELESS SERVICING MANUAL
response in the circuits
coupling the detector to the
deflector plates . When it is
found it can be overcome by
increasing the time-base
frequency, but as this may
introduce another form of
distortion it is better to
improve the circuits
responsible.
The response in the oscil-
r- Tu j j Ml c u loscope circuits must be
Fig. 123 ; The dotted curve illustrates how the • ^ • j j ^ 4 .U
trace of a resonance curve can be distorted by an maintained dOWn tO tne
inadequate low-frequency response in the oscil- fundamental time-base fre-
loscope amplifier , ,
quency — 25 c/s or 12*5 c/s,
whichever be adopted . This means that the oscillocope amplifier
must be a good one ; large capacity coupling condensers must
be used and particular attention paid to grid bias circuits.
Because of this effect it is rarely satisfactory to use the A.F.
amplifier of the receiver and the input to the oscilloscope should
be taken directly from the detector. As resistance-capacity
coupling is used between the detector and the first A.F. stage
in most modern sets, it is rarely necessary to disconnect any-
thing in the receiver and it suffices to clip the oscilloscope input
lead on to the detector output.
In some sets, however, choke or transformer coupling is used
immediately after the detector and it is then imperative to dis-
connect this component if a very distorted trace is to be avoided.
When using this equipment A.V.C. should be rendered in-
operative and this is usually easily done by short-circuiting one
of the A.V.C. filter condensers by a lead with a crocodile clip at
each end. With these precautions no difficulty should be
experienced in obtaining a good trace.
Little need be said regarding the actual circuit adjustments
when using cathode-ray equipment, for it is obvious that the
various trimmers are adjusted for the desired shape of resonance
curve combined with the maximum height of trace. It is thus
quite easy to trim an amplifier with overcoupled tuned circuits,
for the precise effect of every adjustment is clearly shown. In
no other way can such circuits be so reasily adjusted.
There are, of course, many applications of the oscilloscope
beyond those mentioned here, but most of them belong more to
measurement than to general servicing. Their discussion in
this book would consequently be out of place.
276
APPENDIX I
Common Circuit Constants
Condensers
Position
Circuit
Capacity
Bias Resistance by-pass
R.F. or I.F.
o-i ftF
A.F.
25-250 /xF
(old sets)
1-4 /xF
Grid Circuit Decoupling
R.F. or I.F.
0*05-01 /xF
A.F.
0*5-1 /xF
Screen-grid Circuit
R.F. or I.F.
0*1 /xF
Decoupling
AF.
0*5-4
Anode Circuit Decoupling
R.F. or I.F.
0*1-1 /ixF
A.F.
0*5-8 ^F
Inter-stage Coupling
R.F.
100-1,000 /x/xF
Grid Detector
100-300 /x/xF
A.F.
0*01-0*1 jLxF
R.C.-fed Transformer
A.F.
0*1-2 /xF
Anode by-pass
Grid Detector
100-2,000 /x/xF
Anode Bend Detector
100-200 /x/xF
Load Resistance by-pass
Diode Detector
100-300 /x^F
R.F. Filter
Diode Detector
100-500 /U/xF
Output Feed
Choke Output
1-4 /xF
Reservoir Condenser
A.C. Sets
4 mF
Universal
4-32 ^F
Smoothing Condensers
A.C. or D.C.
4-8 /xF
Band-Pass Filter Coupling
“ Bottom-End
0*01-0*05 mF
“ Top-End
0*5-2 /x/xF
Tuning
M.W. and L.W. bands
500 /x/xF
All-wave
350 fX/xF
Short-wave
100-200 /LX/hF
Reaction
M.W. and L.W.
100-500 /lx/xF
All-wave
100-300 /Lt/xF
Short-wave
100-200 fX/xF
Earth Lead Isolating
D.C. Sets
1-2 /lxF
Condensers
Universal
0*1 /xF (max.)
Aerial Lead Isolating
D.C. Sets
0*001-0*1 /xF
Condensers
Universal
0*001-0*01 /txF
(max.)
Mains Aerial
—
X 00-200 /x/xF
Mains to Earth
A.C. and Universal
0*001-0*01 fxF
(max.)
D.C.
0*001-4 mF
Valve Heater to Eaith
A.C. sets ; M.W. and
L.W.
0*1 /iF
S.W.
0*01 /xF
277
WIRELESS SERVICING MANUAL
Resistances
Position
Bias Resistance
Grid-circuit Decoupling
Screen-circuit
Decoupling
Anode Circuit
Decoupling
Anode Coupling
Grid Leak
Load Resistance
Volume Control
Tone Control
Grid Stopping (Anti-
Parasitic)
Anode Stopping (Anti
Parasitic)
Circuit
S.G. or R.F. Pentode
A.F. Triode
Output Triode
Output Pentode
Anode Bend Detector
R.F. or I.F.
A.F.
R.F. or I.F.
A.F.
R.F. or I.F.
A.F.
A.F.
Grid Detector
Anode Bend Detector
R.F. oi I.F.
Grid Detector
A.F.
Diode Detector
Bias on V.M. Valve (i)
A.F. Potentiometer
A.F.
R.F.
F.C.
I.F.
A.F.
R.F. or I.F.
A.F.
The D.C. Resistance of Coils and Chokes
Component
Tuning Coil
Transformers
Mains Transformers
Chokes
Speaker Field
Circuit
M.W.
L.W.
I.F.
A.F., Primary
Secondary
Class “ B,*' Primary
Secondary
Output, Primary
Secondary
Primary
H.T. Secondary
R.F.
R.F., S.W.
Smoothing, ist Stage
2nd Stage
Series fed
Shunt fed
278
Value
50-500 Q
500-2,000 n
500-1,000 Q
145-500 O
2.000- 10,000 12
20.000- 500,000 Q
50.000- 250,000 12
100-1,000 12
2.000- 10,000 12
500-10,000 12
5 . 000 - 100,000 12
10.000- 100,000 12
20.000- 50,000 12
0*1-0*25 M Q
1-2 M 12
0*1-5 M 12
0*1-1 M 12
o*i-o*5 M 12
10.000- 20,000 O
0*1-1 M12
0*1-1 M12
20-200 12
20-500 12
100-500 12
1.000- 10,000 12
100-500 12
50-100 Q
Resistance
i~5 O
5-50 O
5-100 12
500-2,000 12
2.000- 20,000 12
500-2,000 12
100-500 12
200-500 12
0*05-20 12
20-150 O
100-500 12
200-1,000 12
20-100 O
100-300 Q
200-1,000 12
1.000- 2,500 12
2,500-7,500 O
STANDARD SYMBOLS
Standard Symbols
Capacity. Symbol C. Unit, farad.
/nF or mfd. microfarad =* one-millionth of a farad.
fifxF or mmfd. *= micro-microfarad = one-millionth of a
microfarad, also pF (often pronounced ‘‘ puff ” !) = pico-farad
= (Some Continental components have capacity ex-
pressed in centimetres, i cm. = i - 1 /u/uF.)
Inductance. Symbol L. Unit, henry.
H = henry.
mH millihenry = one-thousandth of a henry.
= microhenry = one-millionth of a henry.
Resistance. Symbol R. Unit, ohm.
O = ohm.
kl2 = kilo-ohm = one thousand ohms.
Mo = megohm = one million ohms.
Frequency. Symbol f. Unit, cycles per second
or c/s = cycles per second,
kc/s = kilocycles per second (i.e. thousands of c/s).
Mc/s=* megacycles per second (i.e. millions of c/s).
tti «= 2 Trf 6*28 f.
Voltage, Current and Power
V — volt.
mV millivolt = one-thousandth of a volt
/iV *= microvolt «= one-millionth of a volt.
A ■= ampere.
mA «= milliampere = one-thousandth of an ampere.
fiA = microampere «= one-millionth of an ampere.
W *= watts »= product of volts and amperes (in a resistive
circuit).
mW = Milliwatt =- one-thousandth of a watt.
fxW = Microwatt == one-millionth of a watt.
Volt-Amp «== product of volts and amperes (only equals power
in a resistive circuit).
kVA (usually KVA) = kilo-volt-amperes.
Various
Q = a>L/R = ratio of reactance to resistance of a coil . It is a
measure of its ** goodness*’ and is sometimes called
the magnification and denoted by the letter m.
Rd ac a>* L"/R = <oLQ = dynamic resistance of a tuned circuit.
DB or db = decibel. It is defined as ten times the logarithm
of the ratio of two powers, or twenty times the
logarithm of the ratio of two voltages or currents,
both voltages or currents appearing in the same
value of resistance. It is also commonly used to
N express voltage amplification by taking twenty
times the ratio of the output to input voltages
irrespective of the circuit impedances. While
this usage does not conform to the proper
definition given above, it is common practice.
279
WIRELESS SERVICING MANUAL
Valves
/iorm = amplification factor = dVa/dVg,
Ra internal A.C. resistance (ohms) == dVa/dIa,
g = mutual conductance (A/V) = dIa/dVg =» /x/Ra.,
where dVa, dVg, dia are a change of anode volts, grid volts, and
anode current respectively, and current is expressed in amperes.
Mutual conductance is usually measured in milliamperes per volt'
(written mA/V) or microamperes per volt (/xA/V). It is sometimes
expressed in the conductance term mho (the mho is the reciprocal of
the ohm), i micromho is equal to i /xA/V.
la = anode current. Va — anode voltage.
Ig = grid current. Vg = grid voltage,
lag *= screen current. Vsg = screen voltage.
If = filament current. Vf = filament voltage.
Ic = cathode current. Vc = cathode voltage.
Reactance and Impedance. Units, ohms.
X = reactance.
XI == inductive reactance.
Xc = capacitive reactance.
Z = impedance » combination of resistance and reactance.
Y — i/Z =- admittance.
G == i/R ~ conductance.
S (sometimes B) == i/X — susceptance.
Amplifiers
Class “ A.” Valves operated with a grid voltage about half-way
between zero and the value needed to give
anode current cut-off. No grid current flows
during operation.
Class “ AB ” Push-pull stage with bias intermediate between
the Class ** A ’’ and Class “ B conditions.
Class “ B ” Push pull stage with grid bias near the current
cut-off point.
(Subscripts “ x ” and “ i are commonly used with the above
designations ; ‘S '' indicates that grid current is not permitted,
while “ * *’ denotes that the stage is driven into grid current.)
Q.P.P. A Class *' ABx or Class ‘ Bx stage.
Useful Formulae
Resonance frequency of any low-rcoietance circuit
f = 1/6-28 -/LC (L in H, C in F)
f = 1,000/6-28 \/LC (L in /xH, C in /x/:xF)
Wavelength A =* 1,885 VLC (L in /tH, C in /iF)
and f = 3 X 10*/ A (f in c/s, A in metres)
f = 300/A (f in Mc/s, A in metres)
Ohm’s Law V = IR, I = V/R, R - V/I
in volts, amperes and ohms.
280
USEFUL FORMULAE
Reactance of a coil
Xl - a>L =* 6-a8 fL
(f and L in c/s and H
or Mc/s and /xH)
Reactance of a condenser
Xc — i/ojC « — 1/6-28 fC
, (f and C in c/s and F)
Xc = — io®/6*28fC (f and C in Mc/s and ft/xF)
Resistances in Series
Rt = Rt 4“ R2 4"
Resistances in Parallel
i/Rt = i/Ri 4- 1/R2 4-
Sum of Reactance and Resistance
Z = VR" 4-
Condensers in Series
i/Ct - i/Ct 4- 1/C2 4-
Condensers in Parallel
Ct = Ct 4* C2 4“
The peak value of alternating voltage or current (of sine waveform)
is V 2 = 1*414 times the R.M.S. value. In computing power or
volt-amps. R.M.S. values must always be taken. Unless a voltage
or current is definitely stated to be a peak value, an R.M.S. value
should always be assumed. Thus in dealing with A.C. circuits,
the statement that a valve heater, for instance, must be operated at
4 volts means 4 volts R.M.S., not 4 volts peak, for 4 volts peak is
only 2*828 volts R.M.S.
The above ratio for peak to R.M.S. values is true only for sine
waves, and is different for other waveforms. Thus, take the case
of a modulated carrier. The carrier is usually expressed in R.M.S.
volts and the modulation depth as a percentage. A carrier of 1 volt
when unmodulated has a peak value of 1*414 volts and during
100 per cent, modulation it rises to 2*828 volts peak.
Valve Operating Conditions
Given the operating conditions for one anode voltage Vj, to
determine the correct conditions for a different voltage V2 —
(a) Multiply all voltages by the voltage factor (V2/V1)
(b) Multiply all currents by the current factor (Vj/Vj)®^^
(r) Multiply mutual conductance by the factor (V 2/ V 1) '
(d) Multiply the power output by the factor (V2/V1)®/*
(This rule holds approximately for voltage changes not exceeding
2-5 : I.)
J 281
WIRELESS SERVICING MANUAL
Superheterodyne Ganging
rsiHHi
SIGNAL CIRCUIT OSCILLATOR CIRCUIT
Co ~ Signal circuit stray capacity including trimmer.
Lj = Signal circuit inductance.
Ca = Oscillator padding capacity.
C3 = Oscillator circuit stray capacity including trimmer.
C4 = Any shunt capacity across the oscillator variable
condenser.
Lg = Oscillator circuit inductance.
Let fmin and fmax be the minimum and maximum signal frequencies
to which it is required to tune and Cmin and Cmaz be the minimum
and maximum capacities of the tuning condenser.
Then Cmex f*in*x
c. = Cmia ^ (1)
Pmu. ~ ’
j ^ IQ* ^ / \
W^nux (Co H“ Omla) OJ^atln “H Lutftx)
Next determine the three frequencies at which the ganging is to
be perfect (i.e., the ganging frequencies) as follows : —
Let f', f" and £"' be these frequencies, and enumerate
F
fmax ““ fmio
(3)
f.
** fmla + F/4
(4)
f.
*■ fmax — F/4
(s)
Let fi
=* intermediate frequency.
SUPERHETERODYNE GANGING
Calculate
Yi
» F
1-25 —
I fs + fi
4 * fa 4 - fi
(6)
10-66
fa + fi . _ .
fmln + fi
Ya
= F
^ 1 ^ fs + fi
1 4 • f 7 +;i.
,, fs + ft
+ 3-33
( 7 )
Ya
3 =
F
i6 ' fj
F _i. Ys — yi
+ f i ^ 3
(8;
Then f'
= fmln "f* Yi
( 9 )
f"
= fmln -h fmai
2
— Yt
(lO)
— fmax
— Ya
(II)
The maximum ganging error is given by
Jf = f, /fa ~ f? + f 3 _.
(is)
Now evaluate the oscillator frequencies corresponding to the
signal frequencies : —
fo' « f' + fl
« f" 4 - f,
f/" - f"' 4 - f,
1
J
|(i3)
Writing oj' for 6-28f', <vo' for 6•28fo^ etc.
Evaluate
a « S =-
p = i/a>"» « =
y «= i/o)'* 7 } **
l/a.o"'‘ 1
1/0*0 ">
1/0*0'" J
|(i4)
17 — €
(15)
203
WIRELESS SERVICING MANUAL
The next step is to evaluate the capacities in the tuning condenser
corresponding to the ganging frequencies, and add on C4.
C'" = io« - Co + Cj I
C" = f io‘ - Co + C4 - (16)
*
C' = io‘ - Co + C4 )
Then C4 = C" (6 C'" + C') - C' C'" (i + ff) (17)
C'" + ec' - C" (i + f)
L, = (Ca + C')(C 4 + C'")(’) - 8) (18)
Cj* (C' - C ”0
Cc = J _ C, C'" (19)
L, C, + C'"
In these equations frequency is in Mc/s, inductance in /liH and
capacity in fi/xF.
Note, — When evaluating (17) slide-rule accuracy is not sufficient,
for the denominator involves the difference of two quantities which
are nearly equal.
As an example, suppose fmm = 0 55 Mc/s, fmax — - I ‘ 5 Mc/s,
fl = 0*45 Mc/s, Cratn =15 ft/uF and Cmax == 500 /X/LcF.
From (i) Co = 60-2
(2) Li - 149*5
(3) F =0*95 Mc/s
(4) fa 0*7875 Mc/s
(5) ^3 ~ 1*2625 Mc/s
(6) Yi ^ 0 052 Mc/s
(7) Ya 0*0822 Mc/s
(8) ya - 0*0556 Mc/s
(9) f' 0*602 Mc/s I
(10) f" 0*969 Mc/s >the ganging frequencies.
(11) -- 1*418 Mc/sJ
(12) Jf ^ 0*0033 Mc/s 3’3 kc/s-> the maximum
284
SHORT-'WAVE AERIALS
fC'" •=
From (16) jC" =
Ic' =
(« 7 ) C. =»
(18) Lj —
(19) C3 =
24*3 fifiF,
123*8 fifiF.
407*8 (JLflF.
767 /j,fxF.
65 /xH.
88*45
If C4 = 10 ft/xF. then
rc"' =
From(i6)JC" =
Ic' =
34*3 /x/iF.
133*8 fLflF.
417*8 fifiF.
Short-Wave Aerials and Feeders
Length of half-wave aerial in metres=o*475A=*i42*5/f
in feet =i*56A ==468/f
Length of quarter- wave matching section is one-half of that of
a half-wave aerial.
A reflector should be 0*485 A long and mounted 0*25 A behind the
aerial.
A = wavelength (metres) f = frequency (Mc/s)
Impedance of twin wire feeder or A/4 matching section
Z
=276 log
wire spacmg
Impedance of concentric feeder
Z =118 loff diameter of outer conductor
outside diameter of inner conductor
A quarter-wave matching section must have an impedance
Z **^/ZaZf
where Za is the aerial impedance and Zf is the feeder impedance.
285
WIRELESS SERVICING MANUAL
Copper Wire Tables
Bar* Copper Wire
S.W.G.
1
Diameter
Section
Area
Ohms
per
1,000 yds.
Length
per
ohm
Weight
per
1,000 yds.
Current
Rating at
1,000 amps
per sq. in.
I
in.
aq. in.
yds.
lb.
amps.
lO
0*128
0*02186
1*87
535
148 *8
12*9
12
0*104
0-00849
2*83
353
98*2
8*5
14
0*08
0*00503
4-78
208
58*1
5*0
i2
0*064
0 003 2 2
7*46
13s
37*2
3*2
i8
0*048
0-00180
13*27
53
20*9
I *8
20
0*036
0-00102
23*6
42*4
11*8
1*0
22
0*028
0*00616
39*0
256
7*12
0*61
24
0*022
0*000380
63*2
15*8
4*4
0*38
26
0‘0l8
0*000254
94*3
1
2*94
0*25
28
0*0148
0*000172
139*5
7*i8
1*99
0*17
30
0*0124
0*000121
199
i 5 03
1*40
0*12
32
0*0 108
0 *00009 1
! 262
382
1 *06
0*09
34
0*0092
0*000066
361
; 2*77
0*77
0*07
36
0*0076
0*000045
! X ‘89
0*52
0*05
38
0*006
0*000028
! 849
X‘i8
in.
0*33
oz.
0*03
mA.
40
0*0048
0*000018
1*326
3*35
i8'o
42
0*0040
0*000012
1,910
18*87
2*32
12*6
44
0*0032
0 *000008
2,985
3,899
10*77
1*49
8*0
45
0*0028
0 *000006
6>o
46
0*0024
0*000004
5,307
6*78
0-83
4*5
^2
0*0020
0*000003
7,642
4*71
0*581
3*1
4 «
0*0016
0 *000002
11,941
3*02
0*372
2*0
49
0*0012
0*000001
21.230
1*70
0*209
1*1
Enamelled Copper
; Q
Turns
Weight per
Yards
Ohms per
Ohms
per inch
1,000 yds.
per lb.
1,000 yds.
per lb.
lb.
16
.14*8
37*8
26*4
209*4
5*63
x8
19*7
i 2x3
46*9
37 x *8
17*76
20
26*1
X2'0
83*3
661*3
56*18
22
33*3
7*3
137
1,093
153*51
1 42 x
4*52
22X
1,770
403 -oo
26
i 50*6
3*03
2,645
899 05
28
! 6x*4
2 05
^8
3,914
1,968
30
1 23 ’3 1
1*44
694
5,575
3.985
32
1 h
1*09
915
7,350
6,940
34
1 98
0*832
X,202
10,128
13,174
36
0*543
1,840
14,840
28,272
38
143
0*340
2,810
23,808
72,764
oz.
per oz.
per oz.
40
x8o
3*49
286
11,103
42
2X7
2*43
1"
23,013
44
270
1*56
642
56,226
n
303
357
1*105
0*885
95,404
176,094
47
0*613
1.630
IIHHHiH
368,946
Eureka Wire
286
COPPER WIRE TABLES
Insulated Wire
Single Cotton Covered
Double Cotton Covered
S.W.G.
Turns j Weight per
per inch | 1,000 yds.
;
Yards
per lb. !
Turns
per inch
Weight per^
1,000 yds. 1
Yards
per lb.
.6 j
1
14*1
lb.
38*3
26*1 1
13*3
lb.
38*9
25*6
i8
i 8-3
21 ‘6
46*3
17*3
22*0 j
45*4
20 '
34*1 1
12*24
81*7
21*7
12*6 1
79*2
22 1
29*8 1
7*48
134
263
7*76 I
129
34
37*0
4*57
219
31*3
4*92
203
1
43*5 1
3*21
311
35*7
3 *40
294
28
50*5 i
2*21
452
40*2
2*37
422
30 '
57*5
1*58
634
44*7
1*70 ,
587
32
63*3
1*20
835
50*5
1*32 1
755
34
70-5
0*781
1,280
54*9
0*977 i
1,024
36
00
0^
0*619
1,610
64*1
0*677
1.477
38
100*0
0*392
2,550
71*4
0*437 ,
2,287
40
113*5
0*255
3.910
78*1
0*290 ' 3.456
-
Single Silk Covered
Double Silk Covered
15
37*9
26*4 j
14*7
38*3
26*1
.8 1
20
21*3
46 8
19*6
21 *6
46*3
20 i
26*3
12*0
85*3 1
25*3
12*1
82*5
22
33*3
7*3
137 1
31*8
7*44
134
34
43*1
4*5
1 322 i
40
4*59
218
26
50*6
3*02
j 332 1
47*6
3 08
325
28
60 *4
3*05
00
00
-d-
56*2
2*09
478
30
73*0
1*44
695
67*1
1*48
67s
32
00
1*10
912
75'2
1*13
887
34
93*4
0*80
1.250
85-5
0*82
1,220
36
110
0*551
1.815
102
0*57
1.730
38
133
0*348
2.871
121
0*37
2,760
40
159
02.
3*62
per 02.
276
142
02.
3*88
per 02.
258
43
193
3*58
387
161
2*80
358
44
237
1*67
599
185
1*86
536
45
250
1*33
753
200
1*48
67s
46
378
i*oo
1,000
317
1*15
871
47
31a
0*737
I. 37 S
238
0*845
1.190
287
APPENDIX II
British Colour Codes
Note : The standards tabulated in this section
are those recommended by the Radio Component
Manufacturers* Federation.
Resistance Colour Code
Figure
Colour
Figure
Colour
0
Black
5
Green
1
Brown
6
Blue
2
Red
7
Violet
3
Orange
8
Grey
4
Yellow
9
White
Tolerance
Code
i 5 per cent.
Gold
10 per cent.
Silver
_i: 20 per cent.
No colour
Two distinct systems of marking are used. In the older system
illustrated above, the colour of the body represents the first figure
of the resistance value, and the colour of the tip or end-band represents
the second figure. The colour of the spot or centre-band indicates
the number of ciphers following the first two figures.
In the new method illustrated below there are four bands placed
towards one end of the resistance. The colours of bands i and 2
gives respectively the first and second "figures of the resistance value,
and band 3 indicates the number of ciphers following them. Band 4
represents the resistance tolerance.
Examples :
Old system. Red body, green tip, yellow spot =■ ,250,000 ohms.
New system. Red (i), green (2), yellow (3) -• 250,000 ohms.
288
COLOUR CODES
Colour Codt for Wander Plugs (Batteries)
Highest positive H.T.
Red
2nd highest,, ,,
Yellow
3rd „ „ „
Green
4^h ,, ,, ,,
Blue
L.T. positive
Pink
Negative (— L.T., — H.T., + G.B.)
Black
Maximum G.B. negative
Brown
2nd „ „
Grey
3rd „ „
White
Any additional battery lead is Violet, and any centre-tap White.
Colour Code for Fuses
60 mA
Black
I
Amp.
Dark Blue
100 mA
Grey
I'
►5 Amp.
Light Blue
150 mA
Red
2
Amp.
Purple
250 mA
Brown
3
Amp.
White
500 mA
Yellow
5
Amp.
Black and White
750 mA
Green
Fixed Condenser Markings
Connecting leads for multiple blocks
Highest capacity, positive
2nd highest capacity, „
3rd ,, ,, ,,
4th „ „ „
5th ,, ,, ,,
Principal negative lead
2nd „ „
3rd ,, ,,
Centre lead of voltage doubler
condensers
Red
Yellow
Green
Blue
Violet
Black
Brown
Grey
White
Where two capacities are of the same value, the one of higher
voltage rating has the higher colour in the table.
Common positive junctions are marked +
Common negative junctions are marked —
Series connections are marked
Unconnected sections are marked 8c
Thus
8 + 8 denotes two 8 /zF condensers with common positive terminal
8 8 • ,, ,, ,, ,, ,, negative ,,
8 db 8 „ a series voltage>doubler connection
8 8e 8 „ two isolated 8 fiF condensers
289
WIRELESS SERVICING MANUAL
Mains Transformar Colour Coda
The leads for the high voltage secondary in the case of a Westing-
house metal rectifier or a valve half-wave rectifier are coloured red
and red-yellow.
RECTIFIER
HEATER
' GREEN
GREEN & YELLOW
L GREEN
RED
HIGH
VOLTAGE
SECONDARY
RED & YELLOW
I RED
{ BROWN
BROWN 4 YELLOW
BROWN
BLUE
& YELLOW
BLUE
ADDITIONAL
L.T. WINDING
BLUE
BLACK Sc BROWN
BLACK & RED
BLACK Sc YELLOW
BLACK St GREEN
BLACK
WIRE
American Colour Codes
Resistances. Same as British Code.
Fixed Condensers
The code is similar to that for resistances, the figures represented
by the colours being the same. Capacity is expressed in micro-
microfarads (/i/iF) and three coloured dots appear on the trademark
side of the condenser case. They are read from left to right, the
first dot representing the first figure, the second dot the second
figure, and the third the number of cyphers. Thus, three dots in
order red. black, brown, represents 200 /ti/uF=o ooo2 fiF, and
brown, black, red, represents 1,000 ^^F=o*ooi /xF.
When three significant figures are required to express the capacity,
only the first two dots on the trademark side are coloured, the third
being left blank, and the remainder of the code appears on the
reverse side. The left-hand dot on the reverse side indicates the
third figure and the right-hand dot the number of cyphers. Thus,
a condenser having brown and violet on the trademark side, with
green and brown on the other, would have a capacity of 1,750 fifiF
«bo*ooi 75 /iF.
290
COLOUR CODES
Colour Codo for Battory
Highest positive H.T. (+ B max.)
and highest positive H.T.
3rd
Negative H.T.
Positive L.T.
Negative L.T.
Positive G.B.
Maximum negative G.B.
and negative G.B.
(+ B int.)
(+ B det.)
(- B)
(+ A)
(-A)
(-f C)
( — C max.)
( — C low)
Loudspeaker (high potential lead)
(low
)
Connoctions
Red
Maroon and Red
Maroon
Black with Red tracer
Yellow
Black with Yellow tracer
Green
Black with Green tracer
Black and Green
Brown
Black with Brown tracer
Moving-Coil Loud Speakers
Outside of output transformer primary . .
Green
Inside „ „
>>
99 • •
Brown
Centre- tap on „
>>
99 * •
Red
Outside of „
>>
secondary
White
Inside „ „
»*
99 • •
Maroon
Outside of ist field
winding
Yellow
Inside ,, „
>»
99
Black
Tap on ,,
>>
»
Grey
Outside of and
99
Grey
Inside „ „
>*
99
Blue
Note. — Of the colour codes listed in this Appendix, that for
resistances is almost universally employed. The condenser code is
less widely used. The fuse code is used, but there are exceptions,
particularly with fuses not intended primarily for radio purposes.
The codes for wander plugs, battery connections, mains trans-
formers and loudspeakers are rarely used, and unless one knows that
it is a given manufacturer’s practice to employ them, they should not
be relied upon for identifying leads.
291
APPENDIX Hi
Capacity and Resistance Bridge
A Wireless World Wide-Range Testing Set
T here is no doubt that the best method of measuring capacity
is by means of a bridge. This is a comparison method by which
the unknown condenser is compared with another of known
value, and the accuracy depends upon how closely the relative
values of the two can be determined and upon how accurately the
capacity of the standard is knowm. The basic circuit is shown in
Fig. I, and it is easy to see that if the reactances of Ci and Ca are the
same and Ri equals Ra there will be no current through the tele-
phones. If Ri docs not equal Ra, however, there will be a difference
of potential between the two sides of the phones and, consequently,
an audible note in them. As the slider of the potentiometer is
moved, therefore, the strength of the sound in the phones will
vary and at the exact balance point when Ri equals Ra will disappear.
It is not, of course, necessary for the two capacities to be equal,
but if they are not equal the balance point will be different. The
balance point is connected with the relative values of the condensers
by a very simple relation : Riti/cuCa :: Ra ; i/coCi, Consequently
Ri /Ra = Ci /Ca. If one of the condensers, say Ca, has an accurately
Imown value, and the resistances can be accurately measured each
time a balance has been obtained, the unknown capacity is readily
calculated and is Ci = Ri Ca/Ra. This is naturally inconvenient
for general use, but it is easily possible to provide a scale giving a
direct calibration in capacity, and the operation of measuring capacity
then becomes no more than connecting the condenser to the terminals,
adjus^g the control knob to give the balance, and reading off the
capacity directly from the scale.
In theory it is
possible to measure
all capacities from
zero to infinity in a
single range, but this
is hardly possible in
practice with ordinary
potentiometers, for
the scale would
become extremely
cramped at the ends.
With only two ranges,
however, it is easy to
measure any capacity
between lo fi/xF and
10 ^F.
TTie same bridge
may be used also for
292
CAPAOTY AND RESISTANCE BRIDGE
the measurement of resistance if the standard condenser be replaced
by a standard resistance^ and if the values be suitably chosen the
same scale and calibration hold for both resistance and capacity
measurement. This has been done in the bridge described in this
article, and two ran^s are provided. On one range resistances of lo
ohms to 100,000 ohma can be measured, and capacities of lo /u/liF to
100,000 /xjuF (o* i/uF.), while on the other scale the reading is multiplied
by loo and so gives a range of i,ooo ohms to 10,000,000 ohms (10
M^2) or 1,000 fAfjiF (o-ooi /nF) to io,ooo,oico /nfiF (lo /^F).
The complete circuit diagram of the bridge is shown in Fig. 2,
and it will be seen that the control consists of a 20,000-ohm potentio-
meter R5. The switch S3 changes over the connections for capacity
or resistance measurements, and the switch S2 changes the range.
There are two resistance standards, R3 and R4, of i^ooo ohms and
100,000 ohms respectively, and two capacity standards, C4 and C3,
of 0 001 /LtF and o- 1 fiF respectively. The values of these com-
An under view of the bridge. All components are mounted on the panel which
forms the lid of the box
ponents must be accurately known, and if the bridge is to have •any
pretensions to accuracy and reliability good quality components
having values within ± i per cent, of their ratings must be used.
The total value of the balancing potentiometer R5 has no effect
on the accuracy, and it is only necessary that the ratio of the two halves
be the same in different potentiometers for all positions of the slider
in order to permit a printed scale to be used. This requirement is
met quite well when a uniformly wire- wound potentiotneter i» used,
293
Pif. 2 : The complete circuit diafram of the bridge, ahowing the switching
arrangements
294
CAPACITY AND RESISTANCE BRIDGE
and although the accuracy may suffer slightly through the absence of
individual calibration, this is the inevitable price which must be paid
for avoiding the trouble of calibration.
Constructionally, the components are all secured to a wooden
panel which forms the lid of a shallow box, and to which the scale
printed in Fig. 4* may be pasted. The only non-standard part
employed is the pointer, which is best cut from a strip of brass and
screwed to the control knob by two 8BA screws. Holes must be
drilled and tapped in the knob to receive these screws, and some
care is needed in carrying out this operation, for the bakelite is apt
to crumble somewhat.
A source of alternating current is necessary to energise the bridge.
A.C. mains are not suitable, and the simplest arrangement is to use
a buzzer, connecting the input terminals of the bridge across the
coil of the buzzer. The writer must confess, however, to a dislike
of buzzers on account of their unreliability and their mechanical
noise. Cheap buzzers are very troublesome, being difficult to start
buzzing properly and liable at any moment to cease work. Good
reliable buzzers usually cost as much as a valve oscillator and suffer
from the disadvantage that the direct noise often renders it difficult
to distinguish the exact balance point in the phones.
In the writer’s experience a valve oscillator is the best, since it
is both reliable and silent, and it is is not unduly expensive. The
circuit of a suitable oscillator is given in Fig. 3, and it is recom-
mended that it be built as a separate unit so that it is available
for other purposes, such as energising a bridge of different type,
testing A.F. amplifiers, and modulating an R.F. oscillator. A single
valve is used as a Hartley oscillator with a 3H choke tuned by
a O’Oi /xF condenser Ci. Two dry cells are provided for L.T.
and operate the valve through a lo-ohm resistance Rz. With
new batteries the voltage on the valve does not exceed 2.2 volts,
which is no more than would be obtained with a freshly charged
accumulator applied directly to the filament. Experiment has
shown that the valve will continue to oscillate until the filament
potential falls to about 1.5 volts, so that there is a wide latitude for
falling battery voltage, and an adjustable resistance is unnecessary.
The H.T. supply is provided by two 9 -volt batteries, and as negative
H.T. is connected to positive L.T., the filament battery is effective in
increasing the H.T. supply. With new batteries rather more than 21
volts H.T. is available, and oscillation continues until the H.T. falls
below some 15 volts, so that here, also, ample allowance has been
made for falling battery voltage. An on-off switch has been provided,
and a pilot lamp to give warning that the oscillator is on and so remind
one to switch it off when it is no longer being used.
In order to reduce the load of the bridge on the oscillator and pre-
vent wide variations in frequency and amplitude with different
capacities, a 50,000-ohm resistance, R6, is joined in one of the bridge
input leads. If a buzzer be used for energising the bridge it may
* For Fig. 4 see folded inset facing page 328
295
WIRELESS SERVICING MANUAL
prove necessary to short-
circuit this resistance. A
pair of high-resistance
phones must be used, and
with nothing connected to
the “ X ** terminals the
oscillator note should be
quite loud. The condenser
or resistance to be
measured should be
connected to the “ X **
terminals and the switch
set appropriately to “ C ”
or “ R.” In most cases
the value will be approx-
imately known and the
range switch can then be
set appropriately. The
balance point will be found
to be quite definite,
although complete extinc-
tion of the note may not
always be obtained. The minimum, however, is so definite that it
cannot be mistaken ; it is not obscured by oscillator harmonics.
The reason why a null-point is not always obtained is that no
provision is rmde for balancing out the effect of any resistance in the
condenser being tested nor for balancing out the effect of capacity
in the resistance being measured. Normally, these effects are small
with good components, and the extra complication to the bridge and
its operation is not worth while. If it be found, therefore, that with
a particular component a bad balance is secured, it is possible that,
if a condenser, it has a bad power factor, or if a resistance, it has
a high self-capacity.
output! 1 I lOUTPUT
The oscillator section of the capacity and resistance bridge described In Appendix III
CAPACITY AND RESISTANCE BRIDGE
It must be remembered that the reading of capacity obtained
includes the self-capacity of the bridge and connecting leads. This
can readily be determined by balancing the bridge for capacity with
nothing connected to the “ X terminals, and was about 30 /z/uF
for the original instrument. In order to obtain the true capacity of
a condenser, therefore, this figure must be deducted from the reading
obtained. This correction is unimportant for capacities higher than
1,000 /i/zF
The bridge is arranged to fit over the oscillator and
is held In place by spring clips. A view of the com-
plete unit is on page 7
Even when no attempt at individual calibration is made, and the
printed scale is used, care must be taken to set the pointer correctly.
This may, of course, be done by the end marks which indicate the
limits of travel of the pointer, but for the greatest accuracy it would
be wise to obtain a 1,000 ohms resistance within ± i per cent, (such
resistances are not expensive) and to balance the bridge with this
resistance connected to the “ X ** terminals. Leaving the bridge
balanced, set the pointer to read exactly 1,000.
It may be remarked that no earth connection must be used on any
point of the bridge or oscillator. The condenser or resistance under
test must be disconnected from other gear, otherwise appreciable
«rrora may be introduced.
299
WIRELESS SERVICING MANUAL
WIRELESS WORLD
CAPACITY AND
RESISTANCE BRIDGE
LIST OF PARTS
Bridge
I Wire wound potentiometer, 20,000 ohms non-taper, R5
Reliance “ TW ”
I Resistance, 50,000 ohms, i watt, R6 Bulgin
I Resistance, 1,000 ohms, accuracy i i per cent., 1 watt R3
Bulgin
I Resistance, 100,000 ohms, accuracy ± i per cent., i watt
Bulgin
I Fixed condenser, mica o*i ftF., accuracy it i per cent., C3
Muirhead M7i
1 Fixed condenser, mica o*ooi /iF., accuracy it i per cent., C4
Muirhead M71
2 Switches, DPDT, S2, S3 Bulgin S98
I Knob Bulgin K12
6 Ebonite shrouded terminals, “ phones * (2), “ input ** (2),
Xi,*^ X2 " Belling-Lce B
Wood, wire, screws, etc.
Oscillator
I Valve holder, 4>pin
1 Tapped choke, 3 henries
I Resistance, 100 000 ohms, i watt, Ri
1 Resistance, 10 ohms, R2
2 Fixed condensers, o*oi ftF., Ci, C2
I Switch on /off. Si
I Flush signal lamp with 12mm. 60 mA.
I 3 -volt battery
WB Baseboard type
Varley DP18
Bulgin HW25
Bulgin ARio
Dubilier 670
Bulgin S60T
lib Bulgin D9
Ever Ready, Large
Twin 2-cell 800
2 Grid bias batteries, 9 volts
4 Wander plugs Eelex
2 Ebonite shrouded terminals, “ output Belling-Lee ** B
I Valve, L21 Oaram or Marooni
Wood, wire, tcrewa, etc.
300
APPENDIX IV
Valve Testing Bridge
A simple method of measuring valve constants has been
a long’felt want. The construction of inexpensive gear
for the measurement of mutual conductance is here
described in detail^ and it enables all types of valves other
than diodes to be tested
T he measurement of valve constants normally demands quite
elaborate apparatus, and this is especially the case when a high
degree of accuracy is required. It is because of this that the usual
valve-tester operates merely as an emission tester combined with a
very rough check on mutual conductance. The required voltages
are applied to the valve and the anode current noted ; the grid bias
is then increased by a known amount and the new value of anode
current recorded. The change in anode current divided by the
change in grid voltage gives a figure representing mutual conductance,
while the values of anode current themselves give an indication of
the state of emission.
The results are only of value if it is known what the figures should
be for the particular valve under test. The change of grid bias, for
instance, used in estimating mutual conductance must necessarily
be too large for the valve constants to be the same at both voltages,
for it must never be forgotten that the so-called valve constants are
not constant. They depend upon the construction of the valve, it
is true, but also to a very large degree upon the voltages applied to
the valve. With a triode the amplification factor {fi) usually varies
very little with the opera tii\g conditions, but the anode AC resistance
(Ra) varies enormously. The ratio ft/Ra, which is called the mutual
conductance (g) naturally varies also.
The most accurate
method of measuring the
constants is by means of
a suitable A.C. bridge,
and all three constants
can be measured directly.
The apparatus becomes
quite expensive, however,
when it is required to
determine all three with
accuracy over the wide
ranges of values encoun-
tered in practice. The
A.C. resistance of modern
vdves ranges from 500 mutual
ohms for an output triode conductance brldxe
301
WIRELESS SERVICING MANUAL
The bridge wired up, with t vilve Inserted for test
to lo megohms or more for an R.F. pentode at a high grid bias,
while the amplification factor may be as low as 2*o or as high as
10,000. Mutual conductance covers an equally wide range, from
0-001 mA/V to lo-o mA/V.
Were we concerned only with triodes of ordinary design, it would
be as easy to measure one constant as another, but when extreme
values are encountered this is not always the case. In general, it
is easier to measure mutual conductance than eithe? of the other
constants, and the A.C. resistance is the most troublesome. The
most valuable single constant is mutual conductance, for it reveals
more about the condition of a valve than either amplification factor
or A.C. resistance alone.
There are several methods of measuring mutual conductance,
and the simplest, from the point of view of calibrating the apparatus,
is that shown in Fig. i. The condenser Ci serves merely to prevent
the H.T. supply being short-circuited through the source of alternating
current, and ideally the combination of C2 and R2, which are included
merely to isolate the grid of the valve from the H.T. supply, would
cause no attenuation and no phase shift. Given these conditions,
the input voltage is applied to the grid of the valve directly and to
the anode through Ri. Owing to the reversal of phase which takes
place in the valve, two voltages of opposing phase appear in the anode
circuit, and when they are exactly equal in magnitude no sound can
be heard in the telephones. The magnitude of the voltage which
comes via the valve depends upon the mutual conductance and the
magnitude of the opposing voltage upon the value of Ri.
In practice the value of Ri is varied until silence is obtained in the
telephones, and it can be shown that the mutual conductance in
milliamperes per volt is equal to 1,000 'Ri where Ri is in ohms. In
order to measure mutual conductance over the range of lo-o mA/V
to 0*01 mA/V, therefore, Ri must be variable fiom 100 ohms to
100,000 ohms.
The condenser C2 and resistance R2 will introduce an error if
they cause the voltage actually applied to the grid of the valve to be
302
VALVE TESTING BRIDGE
ippreciably less than that fed to Ri. The error can be made negligibly
small, however, by assigning large enough values to these components.
Of greater importance is the phase shift which they introduce. When
the grid and anode voltages are not in exactly the same phase, no
silent point can be found when adjusting Ri for balance, but only
a minimum. This makes the adjustment difficult and liable to error.
Minimum phase change and the best balance are secured by using
large values for C2 and R2, so that the requirement is the same
as for avoiding attenuation ; it is, however, much more severe.
At first sight it would seem that one could use for these components
the values customarily employed in a resistance-coupled amplifier.
It is, however, often required to take measurements of valves operating
with zero grid bias, for the makers* figures for mutual conductance
are usually quoted for this condition. The valve may then pass grid
current, and if there is a resistance in the grid circuit the bias will
no longer be zero but slightly negative. In order to make the
discrepancy negligible it is necessary to make R2 quite low, and a
value of 2,000 ohms has been found satisfactory. With this low
value of resistance it is necessary to give C2 a capacity of 8 ftF to
maintain the phase change at a satisfactorily low degree.
Using these values a very good balance is secured over the wide
range of o*i-iO‘o mA/V ; although complete silence is not secured,
the balance point is very sharp and definite. For lower values of
mutual conductance the balance deteriorates somewhat, but is
reasonably accurate down to 0*01 mA/V,
WIRELESS SERVICING MANUAL
Since the mutual conductance is proportional to the reciprocal of
resistance, the scale cannot satisfactorily be made direct reading. In
order to obtain a reasonably open scale it would be necessary to have
a tapered resistance with an extraordinarily high degree of taper —
probably too great to manufacture. The scale is consequently
calibrated in resistance values, and to obtain the mutual conductance
it is necessary either to work out the reciprocal or refer to the con-
version curve given in Fig. 3
photographs and drawings clearly show the construction of
the gear which is designed to work with the oscillator, which is
ascribed in Appendix III, for a Capacity and Resistance Bridge.
The complete circuit diagram appears in Fig, 2, and it will be seen
that three variable resistances are employed for balancing. This is
done in order that a high degree of accuracy may be obtained over a
wide range. The resistance Ra has a maximum value of 1,000 ohms,
^.3 ohms, and R4 100,000 ohms. For mutual conductances
higher than i mA/y oiily the 1,000 ohms resistance Rz should be
P ^ being set at zero. The z 0,000 ohms resistance R3
should bt employed for mutual conductances between i mA/V and
304
VALVE TESTING BRIDGE
0*1 mA/V. while the third resistance R4 is for values down to o-oi
mA/V. Whichever resistance is being used, the other two must be
set at zero. It is, of course, possible to employ the three resistances
in the manner of a decade box, setting the high values only at tenths
of the maximum values. This is somewhat more troublesome, but
is slightly more accurate at high values of resistance.
Three valve holders are embodied but any number of additional
ones can be fitted. The filament sockets are all wired in
parallel and connected to the L.T. terminals. For battery valves a
2-volt accumulator should be joined to those, but for A.C. valves
either a 4-volt accumulator or the 4-volt winding of a mains trans-
former. The remaining sockets of the valve holders cannot be
permanently wired, for the connections required depend upon the
particular valve to be tested. A flex lead terminating in a wander
plug is therefore attached to each socket, and a row of five sockets is
provided for them. When not in use the plugs are kept tidy by
being placed in holes drilled in the wooden panel. Only a few of
the plugs are in use at any one, time.
For directly heated valves the filament plug is inserted in the
cathode socket, but for indirectly heated types the cathode plug is
W/RELESS SERVICING MANUAL
Full constructional and wiring deuiis of the bridge. A list of parts It given
on page 307
306
VALVE TESTING BRIDGE
WIRELESS WORLD
VALVE TESTING BRIDGE
LIST OF PARTS
I Valve holder, 5 -pin (without terminals)
Clix Chassis Mounting Type V i
I Valve holder, 7-pin (without terminals)
Clix Chassis Mounting Type V2
1 Valve holder, g-pin (without terminals)
Clix Chassis Mounting Type V2
2 Fixed condensers, 4 /xF. 500 volts DC test, C2, C3
Dubilier B5
I Fixed condenser, o-i /uF. tubular, Ci Dubilier 4513
I Resistance, 50,000 ohms 4 watt, Ri Bulgin HW23
I Resistance, 2,000 ohms 4 watt, R5 Bulgin HWs
I Potentiometer, wire wound, non-taper, 1,000 ohms, R2
Reliance S/Wio
I Potentiometer wire wound, non-taper, 10,000 ohms, R3
S/Wii
i Potentiometer, wire wound, non- taper, 100,000 ohms, R4
Reliance S/W12
3 Knobs Bulgin K12
I I Ebonite shrouded teiminals, input -f- , input — , phones + ,
phones — , grid -f- , grid — , LT + , LT — , HT -f , HT -f i ,
HT — Belling-Lee “ B
17 Wander plugs Eelex
5 Sockets Eelex
Wood, systoflex, wire, screws, etc.
used and the filament plug left free. Whichever valve sockets they
may be, the grid, screen grid, suppressor grid, and anode must always
be joined to the appropriate sockets in the centre by the flex leads.
If the valve has a top cap or side terminal this must be joined to the
appropriate socket by a separate lead terminating in a plug. Any
other pins of normal valves need not be connected, for metallising
and diode anodes can be left free.
Terminals aie provided for the grid bias and H.T. batteries. For
normal tests when the results will be compared with the valve makers*
figures the grid bias terminals will be short-circuited, and for triodes
100 volts H.T. should be used. Actually the anode potential should be
100 volts, and it will be less than the battery voltage by the drop in
the telephones. Strictly speaking, therefore, the battery voltage
should be adjusted so that the voltage measured between the cathode
and anode sockets on the panel is 100 volts with the valve in place
and operative. When power valves are being measured the current
may be quite large and the voltage drop across a pair of phones will
be high. In order to avoid this and to protect the phones it is best
307
WIRELESS SERVICING MANUAL
to use an output transformer with a low primary resistance. Al-
ternatively, the phones can be shunted by a choke of low D.C. resis-
tance ; the inductance need not be high, for 2H or 3H is sufficient.
Measurements of anode current can, of course, be made by connecting
a suitable milliammeter in the positive lead to the high-tension
battery.
Little more need be said about the uses of the instrument or the
method of connecting it up for different valves, for these are obvious.
The question of calibration must receive attention, however, and this
is easily done with the aid of the resistance bridge already referred
to. A sheet of white paper, or, better, ivorine, should be fastened to
the panel to receive the scale, and the two extreme ends of the three
resistances used for balancing joined to the resistance bridge. Set
R3 and R4 to zero and the resistance bridge to 100 ohms. Adjust
Rz for silence, and mark off the scale for 100 ohms. Then set the
bridge to 200 ohms, and repeat. As many intervening values as are
required can be marked off, and when this resistance has been done,
R3 can be tackled. Here both Rz and R4 must be at zero and this
is calibrated in i,ooo-ohm steps with suitable intermediate steps.
Lastly, R4 is calibrated in io,ooo-ohm steps with Rz and R3 at zero.
The process is quite easy, which is fortunate, for it is not possible to
rely on variable resistances being sufficiently consistent for a printed
scale to be used.
308
INDEX
Distortion, Amplitude, 69
— , Detector, 75
— , Frequency, 65
R.F. and 1 F., 79
Earth, 210
Electric deflection, 261
Electromaj^netic deflection, 265
Extension loud speaker, 205
Feed-back, Acoustic, 131, ig6
— from A.F. circuits 61
— I.F. harmonic, 138, 143
— in television apparatus, 257
— modulation by A.h\, 53
Feeders, Aerial, 214
Filters, Crystal, 189
Fly-back, 261, 266
Focus, Television, 264
Formulae, Useful, 280
Frequency-changers, 148
— , Heptode, 152
— , Octode, 152
— , One-valve, 149
— , Triode-heptode, 152
— , Triode-hexode, 152
— , Two- valve, 148
Frequency, Control circuit, 218
— Distortion, 65
— modulated oscillator, 274
— stability, 154, 159
Ganging, Straight Sets, 91
- Defects, 97
— with capacity and inductance
trimmers, loi
— on long waves, 94
— on medium waves, 93
Ganging, Superheterodynes, 115
with capacity and induct-
ance trimmers, 124
— ^ '^dintT system, 117
snaped-piaic condenser, 123
I F. amplifier, 104
Gramophone pick-up, 207, 247
Grid detectors, 75
Harmonics, I.F., 137, 143
— Oscillator, 138
Heater, Circuits, 64
Cathode insulation, 21, 25
Heptode, 152
Hexode, 152
Howl, Threshold, 184
Hum in A.C./D.C. sets, 42
— in A.F. stage, 37
— , Electric pick-up, 38, 39
— , Filament supply, 34
— , Gramophone, 40, 41
— , Grid bias, 34
-, Loud speaker, 31
— , Modulation, 43
, Output stage, 33
— , Pick-up in A.F. transformer,
36
— , Quick tests for, 40
—, in R.F. stage, 39
— , Screen supply, 35
— , Television, 266
— , Valve 37
I.F. amplifier adjustments, 104
— A.V.C. system, 112
— Estimating value of, 146
— harmonic feed-back, 138
— harmonic interference, 137,
— over-coupled circuits, 108
— * pick-up, 142
- variable selectivity, 117
Image interference, 136
Impedance matching, 202, 215
Inductance, Bridge, 5
— trimmers, 101, 124
Instability, Capacity couplings,
56, 58 . .
— Heater circuits, 64
— Impedance coupling, 56
Instability, Magnetic coupling, 57
— R.F. in A.F. circuits, 61
— Sources of, 55
— Stabilising, 59
— - Variation with tuning, 55
insulation. Valve heater-cathode,
21, 25
Interference, Beat, 137
— from sound transmitter, 258
— I.F. harmonic, 137
— I.F. pick-up, 142
INDEX
Interference, Image, 136
— Local, 84
— Man-made static, 84
— Noise, 81
— Oscillator harmonic, 138
— Parasitic oscillation, 82
— Screened lead-in, 87
— Second channel, 136
— Silencing electric motors, 89
— Superheterodyne whistles, 136
— Suppressors, 89
— Thermal agitation, 81
— Thermostats, 89
— Whistles, 129, 136
Leaky coupling condenser, 21
Local-Distance switch, 79
Local interference, 84
Loose connection, 8i
Loud speakers, iqg
— Baffles, 200
— Box resonance, 201
— Centring cones, 199
Loud speakers, Colour code, 288
— Isxtension, 205
— Mains hum in, 31
— Matching to valve, 202
Magnetic coupling. Instability
due to, 57
Mains equipment, 17
— hum, see Hum
— Transformer, Colour code,
290
Man-made static, 84
Matching ganged condensers, 100
— Impedance, 202, 215
— Speaker and valve, 202
Measurements, Capacity, 292
— Continuity, 4
— Current, 13
— Inductance, 5
— Mutual conductance, 27, 301
— Resistance, 292
— Voltage, 9
Metal rectifiers, A.V.C. with, 177
Meter- readings, Interpretation
of, 16
— Tests, A.C. voltages, 17
— Low mains voltage, 16
Meter-readings, Low volts, high
current, 18
— Low' volts, low current, 18
Meters, Constructing, i
— Ohmmeter, 4
— Milliammeter, 3
— Voltmeter, 2
Microphones, 207
Microphony, 25, 131, 196
Milliammeter, 3
— as tuning indicator, 95
— Constructing a multi-range, 3
— Using the, 13
Mixers, 148
Modulation hum, 43
Motor-boating, A.F., 47
— A.V.C., 53
— modulation, 53
~ R F., 53
Motors, Silencing, 89
Muting, 177
Mutual conductance, 27
Negative feed-back, 250
Neutralising, 245
Noise, Hiss, 81
— I.ocal interference, 84
— Loose connections, 81
— Man-made static, 84
— Parasitic oscillation, 82
— - Thermal agitation, 81
— Valve, 81
Octode, 152
Ohmmeter, 4
Oscillation, see Instability
— Parasitic, 82
Oscillators, 154
— Amplitude of, 156
— beat frequency, 187
— coil connections, 156
— frequency stability, 159
— test, 5
Output rnetcr, 7
— transformer, 203
Ozone, 268
Paraphase amplifiers, 238, 261
Parasitic oscillation
— in A.F. stages, 73
INDEX
Parasitic oscillation in oscillators,
8.3
— in push-pull, 242
— in R.F. stages, 82
Pentode, R.C. coupled, 71
“ Plastic,” 260
Pulling, 195
Push-pull amplifiers, 228
— Bias in, 230
— Decoupling in, 231, 239
— Parasitic oscillation, 242
— Phase-splitters, 234
— Resistance-coupled, 232
— Transformer-coupled, 229
Q.A.V.C., 177
Q.P.P., 71
Reactance valve. 218
Reaction, 244
— in S.W. sets, 183
Resistance bridge, 5
— colour code, 288
— testing, 22
Resonance, Box, 201
Resonant aerials, 213, 285
Screen currents, 22
Screened lead-in, 87
Second-channel interference, 136
Selectivity, Variable, 109
Short-wave Aerials, 213, 285
— Conditions, 180
— Converters, 193
— Cross-modulation, 186
— Dead spots, 185
— Receivers, 180
— Sets, Choosing a detector, 182
— Threshold howl, 184
Signal generators, 7
Signal-noise ratio, 209
Silencing electric motors, 89
Sound transmitter. Interference
from, 258
Stabilising receivers, 58
Standard symbols, 279
Superheterodyne, ganging, 115
— Whistles, 136
Switching, Waveband, 23
Synchronisation, 268
Television, 257
— electric deflection, 261
— electromagnetic deflection, 265
— focus, 264
— Intermittent condenser break-
down, 263
— Tube-base sparking, 268
— Voltage supplies, 267
Test oscillators, 5
Thermal agitation, 81
Thermostats, 89
Threshold howl, 184
Tone control, 68, 248
Tools, I
Transformers,
— colour code, 290
— - distortion, 203
- ratio, 202
Transposed feeder, 88
Trimming I.F. amplifiers. 104
Triode-heptode, 152
Triode-hexode, 152
Tuning control, A.F.C., 217
Tweeters, 203
Ultra short waves, 197
Useful formulae, 280
Valve adaptors, 14
’ Bridge, 301
— - Data, 309
— heater-cathode insulation, 21,
25
— Hum, 37
— Mutual conductance, 301
— Noise, 25, 81
— Testing, 301
— Variable-mu, ii
Variable selectivity, 109
Ventilation, i6i
Vibrating condenser vanes, 13 1,
196
Visual circuit alignment, 274
Voltage measurements, 9
Voltmeter, i
— Constructing a multi-range, 2
Waveband switching, 23
Wavemeter, Absorption, 139
Wavetraps, Adjusting, 126
I N D E X
Whistles, A.F., 129
— Filters, 132
— Fixed, 129
— Inter-station, 132
— Microphonic, 13 1
— - Superheterodyne, 136
— Variable intensity, 132
— Variable pitch, 136
Wire tables, 286
Wireless World capacity and
resistance bridge, 292
valve testing bridge, 301
Wobbulators, 274
328
ELECTROLYTIC (WET AND DRY). PAPER. MICA AND
CERAMIC CAPACITORS FOR RADIO RECEIVERS AND
TRANSMITTERS . TELEGRAPHY AND TELEPHONY .
ELECTRONIC EQUIPMENT ^AND INTERFERENCE SUPPRESSION
CAPACITORS
See pa^e Q95, Appendix III
Built-in multiple plugs and
sockets with mechanical coup-
lers instantaneously complete
all electrical circuits and secure
a number of Units into a
rigid assembly without the use
of tools.
AIDS TO BETTER
SERVICING
Designed for the Radio Engineer.
L.A.L. Units cover every need from D.C.
to R.F. The range Includes : Type 15,
C.R.T. unit (6"'tube and self-contained shift
supplles);Type84,TlmeBase(5-200,000cps.
with sweep expansion ; automatic synchro-
nizing ; push-pull deflection ; single or
triggered sweep ; Type 84Y, Amplifier
(gain 25, cathode follower Input, or 600 ;
response level ± I db 10-100,000 cps.,
useful at 0.5 Me. ; push-pull deflection).
5*inch Model BXSO.S, 500
lines, a . 1 .0; Model BX52,
10,000 lines. £1.4.6.
6i-inch Model BX60, 8,500
lines, £1 .2.6; Model BX62.
10,030 lin;;, £1.6.0.
Whenever a new speaker is
required, fit a Truvoxand make
another customer into a firm
friend. Full technical details
on leaflet SH/142 ; a postcard
will bring it to you. Supplies
from your regular wholesaler—
there^s a range of Extension
Speakers too 1
8-inch Model BX80, 8,000
lines. £1.4.0; Model BX82,
10,500 lines, £1 .10.0.
lO-inch Model BX 1 00. 8,000
lines, £1.10.0; Model
BXI02, 10, 500 lines, £1.17.6.
TRUVOX EN8INEERINB CO. LTD.
EXHIBITION GROUNDS, WEMBLEY
(Wembley 1814)
T.X.24
vii
^ lUjlJlXlAJSAJCm
/ ELECYI^IC/tl.
/ YESTIWO
IWSYREMEmW#
The “AVO”
MULTI-RANGE
ELECTRONIC
TESTMETER
This instrament, which
operates with the simplicity
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meter, is an up-to-date
example of current instru-
ment practice. It is a highly
stable thermionic D.C.
Millivoltmeter with circuit
switching, giving 49 ranges
in the following:
D.C. Volts: 2-5 mV. to lO.OOOv. —Input Resisunce 1 1 M megohms.
D.C. Current: 0*25 to i Amp. —150 mV. drop on all ranges.
A.C. Volts: O'lv. to 2,500v. R.M.S. up to I Mc/s.
With external diode probe O-lv. to 250v. up to 200 Mc/s.
A.C. Output Power: 5 mW. to 5 watts in 6 different load resistances
from 5 to 5,000 ohms.
Decibels: — lOdb. to +20 db.
Capacitance: *0001 to 50|iF.
Resistance: 0*2 ohms to 10 megohms.
Insulation: 0*1 megohm to l.0(M megohms.
The thermionic circuit gives deli- The instrument is quickly set up
cate galvanometer sensitivity to for any of the various tests to be
a robust moving-coil movement. undertaken, a single circuit
It is almost impossible to damage selector switch automatically
by overload and is as easy to removing from the circuit any
handle as the standard “ Avo ” voltages or controls which are not
multi range meter. required for the test ni question.
Fully descriptive leaflet available on application
NowAviUabte: The h.r. avo meter., with all the well-known features of the
Universal AvoMeter, and with a sensitivity of 20,000 ohms per volt
So/e Proprietors and Manufacturers
THE AUTOMATIC COIL WIMDER & ELECTRICAL EQUIPMENT CO. LTD
WmCtr Houtt, DouftM StrMt, London, 8.W.1 Telephone: Victoria 3404-9
Which of these is
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127 SHAKESPEARE HOUSE, 17-19 STRATFORD PLACE, LONDON, W.l
IX
AVO
Oscillators
Model 7 and 40 Meters
Resistance and Capacity Bridges
AvoMinors
CONNOISSEUR
Miniature Moving Iron Pick-ups
FURZEHILL
Oscilloscopes
Beat Frequency Oscillators
TAYLOR
Multi-range Meters
Oscilloscopes
Oscillators
Bridges
MEASURING INSTRUMENTS
(PULLINS)
Multi-range Meters
Single Range Meters
Bridges
EDDYSTONE
Communication Receivers
Specialized U.H.F. Components
U.H.F. Communication Apparatus
WELWYN
High-stability Carbon Resistors
Wire-wound Resistors, 2 to 100 watts
LONDEX
Aerial Relays
Power Relays
ROTHERMEL
Crystal Microphones
Crysut Pick-ups
MULLARD
Oscilloscopes
Cathode Ray Units
Bridges
PARTRIDGE
High-Fidelity Output Transformers
Mains Transformers
WILKINS A WRIGHT
Moving Coil Pick-ups
Q.C.C.
Frequency Control Crystals
Resonator Crystals
Calibration Crystals
WODEN
Mains Transformers
L.F. Chokes
Audio Transformers
S. G. BROWN
Moving Coil Headphones
Adjustable Reed Microphones
Lightweighc Headphones
LEXINGTON Moving Coil Pick-ups
ROLA . . . . Loudspeakers
ERIE . . Resistors
CELESTION Loudspeakers
T. C.C Capacitors
DUBILIER . . Capacitors
LABGEAR
Transmitting Inductances
Variable TX Condensers
Crystal Calibrators
WEARITE
Ceramic Wave-change Switches
Receiver Inductances
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14 SOHO STREET. OXFORD STREET. LONDON. W.l.
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XI
RESISTORS • CERAMICONS • Hi-K CERAMICONS • POTENTIOIIETERS
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Telegf*”'* •
XVI
THE OLD
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\
for perfect part replacement
RADIO RESISTOR CO. LTD.
QUADRANT WORKS, CUMBERLAND ROAD, STANMORE, MIDDX
Telephone : WORdsworth 1138-9
2439
xvii
Somebody is designing
another good circuit
When you 8ee a circuit using Mullard valves
you can be sure it is about the best in its class.
Designers who are so wise in their choice of valves
seldom err in other directions.
fcMEEi
Mullard
THE MASTER VALVE
THE MULLARD WIRELESS SERVICE CO. LTD., CENTURY
HOUSE • SHAFTESBURY AVENUE • LONDON W.C.2
MVM60
xviii
A LABORATORY INSTRUMENT
but NOT
Laboratory
Price I
RANGE: iOO Kc/s<60 Mc/s on
fundamentals (up to 120 Mc/s
on Second Harmonic).
ACCURACY:
Guaranteed within‘'±|l%.
ATTEN U ATI O N : Constant
impedance system embodyinf
a matched 75 ohms trans-
mission line.
STRAY FIELD: Less than 3
microvolts at 60 megacycles.
ILLUMINATED DIAL:
Total scale length 30^.
POWER SUPPLY:
110-210-230-250 volts.
DIMENSIONS:
13" X lOi" X 7J" deep.
WEIGHT: 15 lbs.
The “Advance’* Type ‘E’ Signal
Generator places an instrument of
laboratory class within the hnancial
scope of every radio service engineer
and experimenter. The discerning
engineer will appreciate its accuracy
and stability, its exceptionally wide
range which covers all frequencies re-
quired for radio and television receivers,
and its accurate attenuating system
which enables sensitivity measurements
to be made on highly sensitive receivers
up to 60 Mc/s.
Send for fully descriptive pamphlet.
ADVANCE COMPONENTS LTD.
lACK »OAD, SHERNHAll STRKt, WAIIHAMSTOW, lONOON, t.17
Telephone: LARkswood 4366-7-8
X!
Unit, Time Base, Y Deflection Amplifiers and
internal Power Supplies. The two traces are
presented over the full area of a flat screen tube
of 90 mm. internal diameter and operating at
2kv. Signals are normally fed via the Amplifiers,
with provision for input voltage calibration. The
Time Base is designed for repetitive, triggered,
or single stroke operation, and time measurement
is provided by a directly cal. bra ted Shift Control.
For photographic recording the Model 1035 has
provision for the attachment of a Camera,
Model 1428, which may be operated either
manually or by motor drive. A similar camera.
Model 427, is also available for use with Model
339 Oscillograph.
Further details on application to:-^
A. C. CO80OR I^TDm inrSTRtJDlKiWT DKPT., HICiHBlJRV, LOMDOM, NM
CI/So
XX
A Valve Tester which is simple
to operate yet gives the accuracy
of a laboratory instrument
Complete switching of all electrodes is incor-
porated with interlocked filament switch to
prevent short circuit. A differential relay fully
protects the instrument. Valves can be tested
vmder conditions specified in the valve manu-
facturers* lists without reference to charts, etc.
WRITE FOR LEAFLET 217
FOR FULL TECHNICAL DATA
H. C. Atkins Laboratories, Cumberland Road, Kew. RIC. 2950
Alo^
S—T
LABORATORY INSTRUMENTS
FIRST-GRADE MEASURING AND TESTING EQUIPMENT
FOR PHYSICAL AND COMMUNICATIONS LABORATORIES
This Company is producing, on a restricted scale, a range of
specialised Measuring and Test Equipment of the very highest
quality. Special apparatus is also manufactured to customers*
specifications. If you are equipping a Laboratory please
consult us.
C O N ST A NT- V O LT AG E
POWER SUPPLIES
A range of standard models is
available, of various output
voltages and current ratings.
The output voltage is sub-
stantially independent of
Mains voltage or load current,
within the specified limits.
Illustrated is the Model lOI-A,
which provides an output
voltage from 250 to 400 volts
at up to 250 mA, with a
Stabilisation Ratio greater than
100 and an Internal Resistance
less than I ohm.
HIGH-VOLTAGE TEST SETS
These equipments are designed for high-
voltage insulation tests on all types of
components, including capacitors. Tests
may be made either with DC or AC. All
models include a large scale Electrostatic
Voltmeter which is also available, independ-
ently, for the measurement of high voltages.
Illustrated is the Model 301 -A, which covers
the range from 1,000 to 5,000 volts.
BRIDGES. SIGNAL GENERATORS, OSCILLATORS.
OSCILLOSCOPES. REGULATED POWER SUPPLIES.
PRECISION WOUND TRANSFORMERS & CHOKES
PRECISION, QUALITY AND RELIABILITY
ALL-POWER TRANSFORMERS L™
WIMBLEDON, S.W.19
Telephone ; LIBerty 3303
XXll
PRECISION
CORD DRIVES
COMPONENTS
Now available in five types as
illustrated (left to right) Standard,
R/V. Reverse, *• D ” type^^and
** A " type. All one hole fixing.
GANG CONDENSERS
A wide range is now available
In I. 2, 3 or 4 gang types of
various capacities.
JACKSON
BROS ILONDON) LIMITED
KINGSWAY • WADDON • SURREY
ra£W«N£. TEt£ CRAMS- WALFILCO.
CROYDON 27M-5 PHONE. LONDON
Write for Catalogue No. (S.M. I.)
Telephone : Brixton 6506
PARimGE
nANSFORMBKS LTD
12 PKKfORP PUCE, BRIXTON ROAD. S.W.9
nmmi
TRIUKiOim
for LABORATORY USE
and EXPERIMENTAL
WORK
Specially wound to
cuttomert* requirements.
Prompt Delivery.
As supplied to Netionel
Physical Laboratory,
B.B.C., Post Office,
Government Research Es-
tablishments, etc., and the
author of the ** Wireless
Servicing Manual.**
Catalogue free on request
New Partridge Manual
6/- post free
Xxiii
Universally useful The human hands are
universally useful. With the aid of apparatus designed for a particular
job, their adaptability can be infinitely extended. Designed for the
systematic analysis of faults on defective radio sets or other electrical
equipment, the Weston Model E772 Analyser is universally useful in
the hands of an engineer. Its features include high sensitivity — 20,000
ohms per volt on all D.C. ranges — wide range coverage, accuracy,
robust construction and the quality for which Weston instruments are
world famous. Please ask for details.
W T,0 N
I 772
SANCAMO WESTON LTD.. ENFIELD. MIDDX.
T«l«phon»i Enfield 3434 a 1242
NEW "ACRU 24
ILIMiNT
Totally anclosed, air-tight,
crash-proof and virtually
everlasting.
HSATREFLEOlMfi COVER
»» ELECTRIC
SOLDERING
IRON
Heat-reflecting, balanced,
giving perfect control.
COPPER BITS
Instantly detachable and
interchangeable by ro-
tating locking collar.
No screws, no tools
required.
MICA INSUWTED i
*'*"*^\^ feSrEaCOWREDI
-CAM FACE
IMfiAeE RXING raJMlCrSHAKDl
No .cr«r., no fool. 'ISSSilW \ COOUM6W^,V
THE INSTRUMENT IRON WITH UNPARALLELED FEATURES
World Patents Write now for illustrated
Applied for Leaflet giving full details
THE ACRU ELECTRIC TOOL MFG. Co. Ltd., 123 HYDE RD., MANCHESTER 12
XXV
Our experience in the manu-
facture of Electronic Equipment
is extensive, as the following
products show
MICROPHONES
Carbon • Moving Coil
WAVEMETERS
Absorption • Heterodyne
TRANSFORMERS • CHOKES
Power • Audio-frequency • Radio-frequency
LOUD SPEAKERS
High-frequency • High-fidelity • Diffuser
Reflector
TRANSMITTERS
Amplitude Modulated • Frequency Modulated
AMPLIFIERS
Audio-frequency • Intermediate-frequency
Wide-band Radio-frequency
ATTENUATORS
WHITELEY ELECTRICAL RADIO
CO., LTD. MANSFIELD, NOTTS.
[Melton Metallurgical /ABORATO/jJES]
UqHid Silver, Piollneiii and Hold
for metallising CERAMIC,'
MICA, QUARTZ or GLASS
Low Melting-point SOLDER from 70®C.
Non-corrosive liquid FLUX for all electrical
purposes.
Agents for Austra//a ;
A. S. HARRISON & CO. PTY. LTD.
85 CLARENCE STREET. SYDNEY
IPSWICH RD.» TRADING ESTATE
I SLOUGH, BUCKS. Phon«s slough 20992
RAPID. ACCURATE. ^ RELIABLE
The only satisfactory way of
testing all accumulators—
ACID. JELLY & DRY TYPES ^
Tens of thousands already in use
Combined model tests H/T & L/T
QOIXO BATTERY TESTER,
^ Standard Model
^ Write for leaflet U4. IS
TESTOSCOPE
Indicates 2/30 volts and 100/750
volts. A.C. and D.C. Indispensable
to Electrical and Radio Engineers
Write for Interesting booklet U4A
RUNBAKBN- MANCHESTER /
xxvn
IT COSTS SO LITTLE
-AND MEANS SO J^UCH-TO USE
The finest
cored solder In
the world —
FOR RADIO REPAIRS
Ersin Multicore Solder is the
only solder in the world con-
taining three cores of extra-
active non - corrosive Ersin
Flux. The three-core con-
struction of Ersin Multicore
ensures flux continuity and
the correct proportions of flux
to solder, enabling soldering
operations to be carried out
at high speed and eliminating
waste. With the correct
soldering technique, Ersin
Multicore gives a guaranteed
standard of sound precision
joints and avoids ‘dry’ or
H.R. joints.
Prices for Size I Cartons.
Catalogue
Ref. No.
Alloy
Tin-Lead
S.W.G.
C 16014
60/40
14
The Handyman
(Size 2) Carton, for
use in the home,
contains enough
solder for 200 aver-
age joints.
Price : 6d.
C 16018
60/40
18
102 feet
C 14013
40 60
13 1
25 feet
C 14016
1 40/60
16
53 feet
Order from your factor, in case of cifficulty write to :
MULTICORE SOLDERS LTD.
Melller House, Albemarle Street, London, W.l
Tel. : REGent 1411
xxviii
ENGLISH & AMERICAN
TECHNICAL BOOK
SPECIALISTS
We have the Finest Stock of
Radio and Electronic Engineering
Books in the Country
COMPLETE LIST ON APPLICATION
The
MODERN BOOK CO.
19-23 PRAED STREET
LONDON, W.2
Telephone : PAD. 4185
YOU
can become a first-class
RADIO ENGINEER
The real Service Engineer does not rely on guesswork or
luck. In order to tackle every job efficiently, confidently
and quickly, he knows that he must have a thorough
knowledge of modern circuits.
We specialise in teaching radio and allied subjects. Our
method of Home-Study Tuition is an outstanding success.
Hundreds of students owe their progress and improved
earnings solely to our training.
We invite you to write for free details of our Home-Study
Courses in Radio Reception, Servicing, Transmission,
Television, Radio Calculations and Mathematics. (Please
quote ref. : WSMQ).
T. & C. RADIO COLLEGE
King Bdward Ave., Aylesbury, Bucks
xxix
PREMIER RADIO CO
MORRIS & CO. (RADIO) LTD.
A// Post Orders to :
167 LOWER CLAPTON ROAD, LONDON, E.5 Phone: Amherst 4723
Catlers to :
169 FLEET STREET, LONDON, E.C.4 Phone: Central 2833
Our Comprehensive Price List is a necessity to everyone engaged in the Radio
Industry, it contains hundreds of iterm available from Stock. Price 2{d., post paid.
ELECTROLYTIC CONDENSERS.
Capacity
Working
Voltage
Type
Container
Fitting
Dimensions
Price
8mf
500
Ali Tube
Clip
2" X r
3/-
8 x 8mf
500
Cardboard Tube
Clip
2" X ir
7/6
16 X 8mf
450
Ali 'Cube
Clip
4J'' X ir
6/6
16 y 8mf
350
Ali Tulie
Clip
2' X ir
6/6
40mf
200
Ali Tube
Clip
2J'' X ir
5/-
50mf
12
Cardboard Tube
Wire End
ir X jj-'
ir X w'
1/6
25mf
25
Cardboard Tube
Wire End
1/6
12mf
100
Cardboard Tube
Wire Eml
ir X -ft"
1/6
50m f
50
Cardboard Tube
Clip
3J" X IJ"
3/-
250nif
12
Metal Tube
Nil
2r X r
2/6
4nif
12
Cardboard Tul>e
Wire End
ir X jj"
1/-
50mf
50
Ali Tube
Bolt
2 " X r
2/-
16 X 16mf
Cardl)oard Tulie
Clip
3r X ir
7/6
32nif
350
Ali Tube
Clip
3/11
FIRST GRADE OIL FILLED PAPER CONDENSERS, with miniature stand-off
insulators and fixing clips, 2 mfd, 1,000 v.w., 2/6 or 20 /- per dozen; 2 mfd. 600 v.w.,
1/9 each or 14/- jTor dozen ; 1 mfd. 600 v.w., 1/3 each or 12/- per dozen. Super Quality
Oil Filled Tubulars. Insulation as good as Mica ; *1 mfd. 500 v.w., *5 mfd. 350 v.w. Either
type, 9d, each, or 7/6 per dozen.
We can offer from large stocks TUBULAR, MOULDED MICA, SILVER MICA and
CERAMIC CONDENSERS in all capacities. Special price per dozen 4/6, per gross
50/- assorted,
Scrv’icenien’s Parcels of POTENTIOMETERS, 12 assorted sizes suitable for replace-
ments. 10/6 Carbon, 12/- Wire Wound,
ALUMINIUM CHASSIS. Substantially made of bright aluminium, with four sides,
loin. X Sin. x 2Jin.,7/-; 12in. x 9in. X 2Jin,,7;9; 16in. X Sin. X 2 Jin., 8/6 ; 20in. x
Sin. X 2Jin., 10/6 ; 22in. x lOin. x 2Jin., 13/6.
BUILD YOURSELF AN OSCILLOSCOPE. Tev.t LTnit Type 73— consists of a special-
purixise Oscilloscoi^e that requires only rewiring and the addition of a few Condensers
and Resistors to convert into a standard Oscilloscope, input 230v., 50c. A 3J'' C.R. Tube
and 1 S,IJ.220.A, 1 E,B,34, 1 5Z4, 3 S.P.41, 2 A.E.50, are included. Controls are “ Bright-
ness ”, ” V'elocitv ”, ‘‘X Shift ”, ” Y Shift ”, Focus Amplifier “ in/out ”, ” Calibrate ”,
” on/olT/TX ”. Price fi8/8/0d.
Carriage and packing, 20/-. Complete with Metal Case.
MIDGET T.R.F. RADIO KIT. Build your own Receiver. All parts, including
Valves, Speaker and instructions. Nothing else to buy, 4-valve, wavebands 200-557m.
and 700-2,000 metres, 200-250v. A.C./D.C. mains, or A.C. only. Please state which
is required. Size lOin. x 6in. x 6in. £8/0/1 Id. including tax.
MIDGET SUPERHET RADIO KIT. All parts, including Valves, Speaker and
instructions, to build a S-valve Superhet. 16-50m. and 200-257 metres, 200-250v.
A.C./D.C. or A.C. only. Please state which is required. Size lOin. x 6in. x 6in.
Price £9, including tax. An attractive brown bakelite cabinet can be supplied for either
Kit at 27/3.
XXX
XXXI
lYTttfMt ACCURACY Is
Biillers
tkH miSfy
Made in Three
Principal Materials
CUSTOMW
FREQUELEX
I An insulating material of
Low Di-electric Loss, for
Coil Formers, Aerial In-
sulators, Valve Holders,
PERMALEX
A High Permittivity
Material. For the con-
struction of Condensers
of the smallest possible
dimensions.
TEMPLEX
A Condenser material of
medium permittivity. For
the construction of Con-
densers having a constant
capacity of all tempera-
tures.
BULLERS LOW LOSS CERAMICS
i^ULLERS LTD., 6 Laurence Pountney Hill, London. E.C.4
i^'hone : Mansion House 9971 (3 lines). Telegrams : " Bullers, Cannon, London.'*
xxxii
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RADIO
BASIC MATHEMATICS FOR RADIO
STUDENTS
By F. M. Colebrook, B.Sc., D.I.C., A.CGJ.
I0«. 6d. By post I Os. lOd.
FOUNDATIONS OF WIRELESS
(4th Edition. Revised)
By M. G. Scroggie, B.Sc., M.I.E.E.
7s. 6d. By post Is. lOd.
RADIO LABORATORY HANDBOOK
(4th Edition)
By M. G. Scroggie, B.Sc., M.I.E.E,
12s. 6d. By post 12$. lOd.
TELEVISION RECEIVING EQUIPMENT
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By W. T. Cocking, M.I.E.E.
12s. 6d. By post 12$. lid.
MOTORING
KNOW YOUR CAR (3rd Edition)
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LOOK AFTER YOUR CAR : Everyday
Maintenance Simply Explained
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MOTOR CYCLING
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SPEED FROM YOUR MOTOR CYCLE
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ROAD TRANSPORT
DIESEL MAINTENANCE (2nd Edit.
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AUTOMOBILE CHASSIS DESIGN
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MODERN DIESEL (llth Edition .
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Edited by G. Geoffrey Smith, M.B.E,
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